This article is about the technology of power electronics. For the musical genre, see power electronics (music)
An HVDC thyristor valve tower 16.8 m tall in a hall at Baltic Cable AB in Sweden A battery charger is an example of a piece of power electronics. A PCs power supply is an example of a piece of power electronics, whether inside or outside of the cabinet.Power electronics is the application of electronics to the control and conversion of electric power.
The first high-power electronic devices were made using mercury-arc valves. In modern systems, the conversion is performed with semiconductor switching devices such as diodes, thyristors, and power transistors such as the power MOSFET and IGBT. In contrast to electronic systems concerned with the transmission and processing of signals and data, substantial amounts of electrical energy are processed in power electronics. An AC/DC converter (rectifier) is the most typical power electronics device found in many consumer electronic devices, e.g. television sets, personal computers, battery chargers, etc. The power range is typically from tens of watts to several hundred watts. In industry, a common application is the variable speed drive (VSD) that is used to control an induction motor. The power range of VSDs starts from a few hundred watts and ends at tens of megawatts.
The power conversion systems can be classified according to the type of the input and output power:
History
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Power electronics started with the development of the mercury arc rectifier. Invented by Peter Cooper Hewitt in 1902, it was used to convert alternating current (AC) into direct current (DC). From the 1920s on, research continued on applying thyratrons and grid-controlled mercury arc valves to power transmission. Uno Lamm developed a mercury valve with grading electrodes making them suitable for high voltage direct current power transmission. In 1933 selenium rectifiers were invented.[1]
Julius Edgar Lilienfeld proposed the concept of a field-effect transistor in 1926, but it was not possible to actually construct a working device at that time.[2] In 1947, the bipolar point-contact transistor was invented by Walter H. Brattain and John Bardeen under the direction of William Shockley at Bell Labs. In 1948 Shockley's invention of the bipolar junction transistor (BJT) improved the stability and performance of transistors, and reduced costs. By the 1950s, higher power semiconductor diodes became available and started replacing vacuum tubes. In 1956, the silicon controlled rectifier (SCR) was introduced by General Electric, greatly increasing the range of power electronics applications.[3] By the 1960s, the improved switching speed of bipolar junction transistors had allowed for high frequency DC/DC converters.
R. D. Middlebrook made important contributions to power electronics. In 1970, he founded the Power Electronics Group at Caltech.[4] He developed the state-space averaging method of analysis and other tools crucial to modern power electronics design.[5]
Power MOSFET
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A breakthrough in power electronics came with the invention of the MOSFET (metal–oxide–semiconductor field-effect transistor) by Mohamed Atalla and Dawon Kahng at Bell Labs in 1959. Generations of MOSFET transistors enabled power designers to achieve performance and density levels not possible with bipolar transistors.[6] Due to improvements in MOSFET technology (initially used to produce integrated circuits), the power MOSFET became available in the 1970s.
In 1969, Hitachi introduced the first vertical power MOSFET,[7] which would later be known as the VMOS (V-groove MOSFET).[8] From 1974, Yamaha, JVC, Pioneer Corporation, Sony and Toshiba began manufacturing audio amplifiers with power MOSFETs.[9] International Rectifier introduced a 25 A, 400 V power MOSFET in 1978.[10] This device allows operation at higher frequencies than a bipolar transistor, but is limited to low voltage applications.
The power MOSFET is the most common power device in the world, due to its low gate drive power, fast switching speed,[11] easy advanced paralleling capability,[11][12] wide bandwidth, ruggedness, easy drive, simple biasing, ease of application, and ease of repair.[12] It has a wide range of power electronic applications, such as portable information appliances, power integrated circuits, cell phones, notebook computers, and the communications infrastructure that enables the Internet.[13]
In 1982, the insulated-gate bipolar transistor (IGBT) was introduced. It became widely available in the 1990s. This component has the power handling capability of the bipolar transistor and the advantages of the isolated gate drive of the power MOSFET.
Devices
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The capabilities and economy of power electronics system are determined by the active devices that are available. Their characteristics and limitations are a key element in the design of power electronics systems. Formerly, the mercury arc valve, the high-vacuum and gas-filled diode thermionic rectifiers, and triggered devices such as the thyratron and ignitron were widely used in power electronics. As the ratings of solid-state devices improved in both voltage and current-handling capacity, vacuum devices have been nearly entirely replaced by solid-state devices.
Power electronic devices may be used as switches, or as amplifiers.[14] An ideal switch is either open or closed and so dissipates no power; it withstands an applied voltage and passes no current or passes any amount of current with no voltage drop. Semiconductor devices used as switches can approximate this ideal property and so most power electronic applications rely on switching devices on and off, which makes systems very efficient as very little power is wasted in the switch. By contrast, in the case of the amplifier, the current through the device varies continuously according to a controlled input. The voltage and current at the device terminals follow a load line, and the power dissipation inside the device is large compared with the power delivered to the load.
Several attributes dictate how devices are used. Devices such as diodes conduct when a forward voltage is applied and have no external control of the start of conduction. Power devices such as silicon controlled rectifiers and thyristors (as well as the mercury valve and thyratron) allow control of the start of conduction but rely on periodic reversal of current flow to turn them off. Devices such as gate turn-off thyristors, BJT and MOSFET transistors provide full switching control and can be turned on or off without regard to the current flow through them. Transistor devices also allow proportional amplification, but this is rarely used for systems rated more than a few hundred watts. The control input characteristics of a device also significantly affect design; sometimes, the control input is at a very high voltage with respect to ground and must be driven by an isolated source.
As efficiency is at a premium in a power electronic converter, the losses generated by a power electronic device should be as low as possible.
Devices vary in switching speed. Some diodes and thyristors are suited for relatively slow speed and are useful for power frequency switching and control; certain thyristors are useful at a few kilohertz. Devices such as MOSFETS and BJTs can switch at tens of kilohertz up to a few megahertz in power applications, but with decreasing power levels. Vacuum tube devices dominate high power (hundreds of kilowatts) at very high frequency (hundreds or thousands of megahertz) applications. Faster switching devices minimize energy lost in the transitions from on to off and back but may create problems with radiated electromagnetic interference. Gate drive (or equivalent) circuits must be designed to supply sufficient drive current to achieve the full switching speed possible with a device. A device without sufficient drive to switch rapidly may be destroyed by excess heating.
Practical devices have a non-zero voltage drop and dissipate power when on, and take some time to pass through an active region until they reach the "on" or "off" state. These losses are a significant part of the total lost power in a converter.
Power handling and dissipation of devices is also critical factor in design. Power electronic devices may have to dissipate tens or hundreds of watts of waste heat, even switching as efficiently as possible between conducting and non-conducting states. In the switching mode, the power controlled is much larger than the power dissipated in the switch. The forward voltage drop in the conducting state translates into heat that must be dissipated. High power semiconductors require specialized heat sinks or active cooling systems to manage their junction temperature; exotic semiconductors such as silicon carbide have an advantage over straight silicon in this respect, and germanium, once the main-stay of solid-state electronics is now little used due to its unfavorable high-temperature properties.
Semiconductor devices exist with ratings up to a few kilovolts in a single device. Where very high voltage must be controlled, multiple devices must be used in series, with networks to equalize voltage across all devices. Again, switching speed is a critical factor since the slowest-switching device will have to withstand a disproportionate share of the overall voltage. Mercury valves were once available with ratings to 100 kV in a single unit, simplifying their application in HVDC systems.
The current rating of a semiconductor device is limited by the heat generated within the dies and the heat developed in the resistance of the interconnecting leads. Semiconductor devices must be designed so that current is evenly distributed within the device across its internal junctions (or channels); once a "hot spot" develops, breakdown effects can rapidly destroy the device. Certain SCRs are available with current ratings to 3000 amperes in a single unit.
DC/AC converters (inverters)
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DC to AC converters produce an AC output waveform from a DC source. Applications include adjustable speed drives (ASD), uninterruptible power supplies (UPS), Flexible AC transmission systems (FACTS), voltage compensators, and photovoltaic inverters. Topologies for these converters can be separated into two distinct categories: voltage source inverters and current source inverters. Voltage source inverters (VSIs) are named so because the independently controlled output is a voltage waveform. Similarly, current source inverters (CSIs) are distinct in that the controlled AC output is a current waveform.
DC to AC power conversion is the result of power switching devices, which are commonly fully controllable semiconductor power switches. The output waveforms are therefore made up of discrete values, producing fast transitions rather than smooth ones. For some applications, even a rough approximation of the sinusoidal waveform of AC power is adequate. Where a near sinusoidal waveform is required, the switching devices are operated much faster than the desired output frequency, and the time they spend in either state is controlled so the averaged output is nearly sinusoidal. Common modulation techniques include the carrier-based technique, or Pulse-width modulation, space-vector technique, and the selective-harmonic technique.[15]
Voltage source inverters have practical uses in both single-phase and three-phase applications. Single-phase VSIs utilize half-bridge and full-bridge configurations, and are widely used for power supplies, single-phase UPSs, and elaborate high-power topologies when used in multicell configurations. Three-phase VSIs are used in applications that require sinusoidal voltage waveforms, such as ASDs, UPSs, and some types of FACTS devices such as the STATCOM. They are also used in applications where arbitrary voltages are required, as in the case of active power filters and voltage compensators.[15]
Current source inverters are used to produce an AC output current from a DC current supply. This type of inverter is practical for three-phase applications in which high-quality voltage waveforms are required.
A relatively new class of inverters, called multilevel inverters, has gained widespread interest. The normal operation of CSIs and VSIs can be classified as two-level inverters, due to the fact that power switches connect to either the positive or to the negative DC bus. If more than two voltage levels were available to the inverter output terminals, the AC output could better approximate a sine wave. It is for this reason that multilevel inverters, although more complex and costly, offer higher performance.[16]
Each inverter type differs in the DC links used, and in whether or not they require freewheeling diodes. Either can be made to operate in square-wave or pulse-width modulation (PWM) mode, depending on its intended usage. Square-wave mode offers simplicity, while PWM can be implemented in several different ways and produces higher quality waveforms.[15]
Voltage Source Inverters (VSI) feed the output inverter section from an approximately constant-voltage source.[15]
The desired quality of the current output waveform determines which modulation technique needs to be selected for a given application. The output of a VSI is composed of discrete values. In order to obtain a smooth current waveform, the loads need to be inductive at the select harmonic frequencies. Without some sort of inductive filtering between the source and load, a capacitive load will cause the load to receive a choppy current waveform, with large and frequent current spikes.[15]
There are three main types of VSIs:
Single-phase half-bridge inverter
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Figure 8: The AC input for an ASD FIGURE 9: Single-phase half-bridge voltage source inverterThe single-phase voltage source half-bridge inverters are meant for lower voltage applications and are commonly used in power supplies.[15] Figure 9 shows the circuit schematic of this inverter.
Low-order current harmonics get injected back to the source voltage by the operation of the inverter. This means that two large capacitors are needed for filtering purposes in this design.[15] As Figure 9 illustrates, only one switch can be on at a time in each leg of the inverter. If both switches in a leg were on at the same time, the DC source would be shorted out.
Inverters can use several modulation techniques to control their switching schemes. The carrier-based PWM technique compares the AC output waveform, vc, to a carrier voltage signal, vΔ. When vc is greater than vΔ, S+ is on, and when vc is less than vΔ, S− is on. When the AC output is at frequency fc with its amplitude at vc, and the triangular carrier signal is at frequency fΔ with its amplitude at vΔ, the PWM becomes a special sinusoidal case of the carrier based PWM.[15] This case is dubbed sinusoidal pulse-width modulation (SPWM).For this, the modulation index, or amplitude-modulation ratio, is defined as ma = vc/v∆ .
The normalized carrier frequency, or frequency-modulation ratio, is calculated using the equation mf = f∆/fc .[17]
If the over-modulation region, ma, exceeds one, a higher fundamental AC output voltage will be observed, but at the cost of saturation. For SPWM, the harmonics of the output waveform are at well-defined frequencies and amplitudes. This simplifies the design of the filtering components needed for the low-order current harmonic injection from the operation of the inverter. The maximum output amplitude in this mode of operation is half of the source voltage. If the maximum output amplitude, ma, exceeds 3.24, the output waveform of the inverter becomes a square wave.[15]
As was true for Pulse-Width Modulation (PWM), both switches in a leg for square wave modulation cannot be turned on at the same time, as this would cause a short across the voltage source. The switching scheme requires that both S+ and S− be on for a half cycle of the AC output period.[15] The fundamental AC output amplitude is equal to vo1 = vaN = 2vi/π .
Its harmonics have an amplitude of voh = vo1/h.
Therefore, the AC output voltage is not controlled by the inverter, but rather by the magnitude of the DC input voltage of the inverter.[15]
Using selective harmonic elimination (SHE) as a modulation technique allows the switching of the inverter to selectively eliminate intrinsic harmonics. The fundamental component of the AC output voltage can also be adjusted within a desirable range. Since the AC output voltage obtained from this modulation technique has odd half and odd quarter-wave symmetry, even harmonics do not exist.[15] Any undesirable odd (N-1) intrinsic harmonics from the output waveform can be eliminated.
Single-phase full-bridge inverter
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FIGURE 3: Single-phase voltage source full-bridge inverter FIGURE 4: Carrier and modulating signals for the bipolar pulsewidth modulation techniqueThe full-bridge inverter is similar to the half bridge-inverter, but it has an additional leg to connect the neutral point to the load.[15] Figure 3 shows the circuit schematic of the single-phase voltage source full-bridge inverter.
To avoid shorting out the voltage source, S1+, and S1− cannot be on at the same time, and S2+ and S2− also cannot be on at the same time. Any modulating technique used for the full-bridge configuration should have either the top or the bottom switch of each leg on at any given time. Due to the extra leg, the maximum amplitude of the output waveform is Vi, and is twice as large as the maximum achievable output amplitude for the half-bridge configuration.[15]
States 1 and 2 from Table 2 are used to generate the AC output voltage with bipolar SPWM. The AC output voltage can take on only two values, either Vi or −Vi. To generate these same states using a half-bridge configuration, a carrier based technique can be used. S+ being on for the half-bridge corresponds to S1+ and S2− being on for the full-bridge. Similarly, S− being on for the half-bridge corresponds to S1− and S2+ being on for the full bridge. The output voltage for this modulation technique is more or less sinusoidal, with a fundamental component that has an amplitude in the linear region of less than or equal to one[15] vo1 =vab1= vi • ma.
Unlike the bipolar PWM technique, the unipolar approach uses states 1, 2, 3, and 4 from Table 2 to generate its AC output voltage. Therefore, the AC output voltage can take on the values Vi, 0 or −V [1]i. To generate these states, two sinusoidal modulating signals, Vc and −Vc, are needed, as seen in Figure 4.
Vc is used to generate VaN, while –Vc is used to generate VbN. The following relationship is called unipolar carrier-based SPWM vo1 =2 • vaN1= vi • ma.
The phase voltages VaN and VbN are identical, but 180 degrees out of phase with each other. The output voltage is equal to the difference of the two-phase voltages, and do not contain any even harmonics. Therefore, if mf is taken, even the AC output voltage harmonics will appear at normalized odd frequencies, fh. These frequencies are centered on double the value of the normalized carrier frequency. This particular feature allows for smaller filtering components when trying to obtain a higher quality output waveform.[15]
As was the case for the half-bridge SHE, the AC output voltage contains no even harmonics due to its odd half and odd quarter-wave symmetry.[15]
Three-phase voltage source inverter
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FIGURE 5: Three-phase voltage source inverter circuit schematic FIGURE 6: Three-phase square-wave operation a) Switch state S1 b) Switch state S3 c) S1 output d) S3 outputSingle-phase VSIs are used primarily for low power range applications, while three-phase VSIs cover both medium and high power range applications.[15] Figure 5 shows the circuit schematic for a three-phase VSI.
Switches in any of the three legs of the inverter cannot be switched off simultaneously due to this resulting in the voltages being dependent on the respective line current's polarity. States 7 and 8 produce zero AC line voltages, which result in AC line currents freewheeling through either the upper or the lower components. However, the line voltages for states 1 through 6 produce an AC line voltage consisting of the discrete values of Vi, 0 or −Vi.[15]
For three-phase SPWM, three modulating signals that are 120 degrees out of phase with one another are used in order to produce out-of-phase load voltages. In order to preserve the PWM features with a single carrier signal, the normalized carrier frequency, mf, needs to be a multiple of three. This keeps the magnitude of the phase voltages identical, but out of phase with each other by 120 degrees.[15] The maximum achievable phase voltage amplitude in the linear region, ma less than or equal to one, is vphase = vi / 2. The maximum achievable line voltage amplitude is Vab1 = vab • √3 / 2
The only way to control the load voltage is by changing the input DC voltage.
Current source inverters
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FIGURE 7: Three-phase current source inverter Figure 8: Synchronized-pulse-width-modulation waveforms for a three-phase current source inverter a) Carrier and modulating Ssgnals b) S1 state c) S3 state d) Output current Figure 9: Space-vector representation in current source invertersCurrent source inverters convert DC current into an AC current waveform. In applications requiring sinusoidal AC waveforms, magnitude, frequency, and phase should all be controlled. CSIs have high changes in current over time, so capacitors are commonly employed on the AC side, while inductors are commonly employed on the DC side.[15] Due to the absence of freewheeling diodes, the power circuit is reduced in size and weight, and tends to be more reliable than VSIs.[16] Although single-phase topologies are possible, three-phase CSIs are more practical.
In its most generalized form, a three-phase CSI employs the same conduction sequence as a six-pulse rectifier. At any time, only one common-cathode switch and one common-anode switch are on.[16]
As a result, line currents take discrete values of –ii, 0 and ii. States are chosen such that a desired waveform is output and only valid states are used. This selection is based on modulating techniques, which include carrier-based PWM, selective harmonic elimination, and space-vector techniques.[15]
Carrier-based techniques used for VSIs can also be implemented for CSIs, resulting in CSI line currents that behave in the same way as VSI line voltages. The digital circuit utilized for modulating signals contains a switching pulse generator, a shorting pulse generator, a shorting pulse distributor, and a switching and shorting pulse combiner. A gating signal is produced based on a carrier current and three modulating signals.[15]
A shorting pulse is added to this signal when no top switches and no bottom switches are gated, causing the RMS currents to be equal in all legs. The same methods are utilized for each phase, however, switching variables are 120 degrees out of phase relative to one another, and the current pulses are shifted by a half-cycle with respect to output currents. If a triangular carrier is used with sinusoidal modulating signals, the CSI is said to be utilizing synchronized-pulse-width-modulation (SPWM). If full over-modulation is used in conjunction with SPWM the inverter is said to be in square-wave operation.[15]
The second CSI modulation category, SHE is also similar to its VSI counterpart. Utilizing the gating signals developed for a VSI and a set of synchronizing sinusoidal current signals, results in symmetrically distributed shorting pulses and, therefore, symmetrical gating patterns. This allows any arbitrary number of harmonics to be eliminated.[15] It also allows control of the fundamental line current through the proper selection of primary switching angles. Optimal switching patterns must have quarter-wave and half-wave symmetry, as well as symmetry about 30 degrees and 150 degrees. Switching patterns are never allowed between 60 degrees and 120 degrees. The current ripple can be further reduced with the use of larger output capacitors, or by increasing the number of switching pulses.[16]
The third category, space-vector-based modulation, generates PWM load line currents that equal load line currents, on average. Valid switching states and time selections are made digitally based on space vector transformation. Modulating signals are represented as a complex vector using a transformation equation. For balanced three-phase sinusoidal signals, this vector becomes a fixed module, which rotates at a frequency, ω. These space vectors are then used to approximate the modulating signal. If the signal is between arbitrary vectors, the vectors are combined with the zero vectors I7, I8, or I9.[15] The following equations are used to ensure that the generated currents and the current vectors are on the average equivalent.
Multilevel inverters
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FIGURE 10: Three-level neutral-clamped inverterA relatively new class called multilevel inverters has gained widespread interest. Normal operation of CSIs and VSIs can be classified as two-level inverters because the power switches connect to either the positive or the negative DC bus.[16] If more than two voltage levels were available to the inverter output terminals, the AC output could better approximate a sine wave.[15] For this reason multilevel inverters, although more complex and costly, offer higher performance.[16] A three-level neutral-clamped inverter is shown in Figure 10.
Control methods for a three-level inverter only allow two switches of the four switches in each leg to simultaneously change conduction states. This allows smooth commutation and avoids shoot through by only selecting valid states.[16] It may also be noted that since the DC bus voltage is shared by at least two power valves, their voltage ratings can be less than a two-level counterpart.
Carrier-based and space-vector modulation techniques are used for multilevel topologies. The methods for these techniques follow those of classic inverters, but with added complexity. Space-vector modulation offers a greater number of fixed voltage vectors to be used in approximating the modulation signal, and therefore allows more effective space vector PWM strategies to be accomplished at the cost of more elaborate algorithms. Due to added complexity and the number of semiconductor devices, multilevel inverters are currently more suitable for high-power high-voltage applications.[16] This technology reduces the harmonics hence improves overall efficiency of the scheme.
AC/AC converters
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Converting AC power to AC power allows control of the voltage, frequency, and phase of the waveform applied to a load from a supplied AC system .[18] The two main categories that can be used to separate the types of converters are whether the frequency of the waveform is changed.[19] AC/AC converter that don't allow the user to modify the frequencies are known as AC Voltage Controllers, or AC Regulators. AC converters that allow the user to change the frequency are simply referred to as frequency converters for AC to AC conversion. Under frequency converters there are three different types of converters that are typically used: cycloconverter, matrix converter, DC link converter (aka AC/DC/AC converter).
AC voltage controller: The purpose of an AC Voltage Controller, or AC Regulator, is to vary the RMS voltage across the load while at a constant frequency.[18] Three control methods that are generally accepted are ON/OFF Control, Phase-Angle Control, and Pulse-Width Modulation AC Chopper Control (PWM AC Chopper Control).[20] All three of these methods can be implemented not only in single-phase circuits, but three-phase circuits as well.
Matrix converters and cycloconverters: Cycloconverters are widely used in industry for ac to ac conversion, because they are able to be used in high-power applications. They are commutated direct frequency converters that are synchronised by a supply line. The cycloconverters output voltage waveforms have complex harmonics with the higher-order harmonics being filtered by the machine inductance. Causing the machine current to have fewer harmonics, while the remaining harmonics causes losses and torque pulsations. Note that in a cycloconverter, unlike other converters, there are no inductors or capacitors, i.e. no storage devices. For this reason, the instantaneous input power and the output power are equal.[21]
when?
] because of the decrease in both size and price of the power electronics switches. The single-phase high frequency ac voltage can be either sinusoidal or trapezoidal. These might be zero voltage intervals for control purpose or zero voltage commutation.With recent device advances, newer forms of cycloconverters are being developed, such as matrix converters. The first change that is first noticed is that matrix converters utilize bi-directional, bipolar switches. A single phase to a single phase matrix converter consists of a matrix of 9 switches connecting the three input phases to the tree output phase. Any input phase and output phase can be connected together at any time without connecting any two switches from the same phase at the same time; otherwise this will cause a short circuit of the input phases. Matrix converters are lighter, more compact and versatile than other converter solutions. As a result, they are able to achieve higher levels of integration, higher temperature operation, broad output frequency and natural bi-directional power flow suitable to regenerate energy back to the utility.
The matrix converters are subdivided into two types: direct and indirect converters. A direct matrix converter with three-phase input and three-phase output, the switches in a matrix converter must be bi-directional, that is, they must be able to block voltages of either polarity and to conduct current in either direction. This switching strategy permits the highest possible output voltage and reduces the reactive line-side current. Therefore, the power flow through the converter is reversible. Because of its commutation problem and complex control keep it from being broadly utilized in industry.
Unlike the direct matrix converters, the indirect matrix converters has the same functionality, but uses separate input and output sections that are connected through a dc link without storage elements. The design includes a four-quadrant current source rectifier and a voltage source inverter. The input section consists of bi-directional bipolar switches. The commutation strategy can be applied by changing the switching state of the input section while the output section is in a freewheeling mode. This commutation algorithm is significantly less complex, and has higher reliability as compared to a conventional direct matrix converter.[22]
DC link converters: DC Link Converters, also referred to as AC/DC/AC converters, convert an AC input to an AC output with the use of a DC link in the middle. Meaning that the power in the converter is converted to DC from AC with the use of a rectifier, and then it is converted back to AC from DC with the use of an inverter. The end result is an output with a lower voltage and variable (higher or lower) frequency.[20] Due to their wide area of application, the AC/DC/AC converters are the most common contemporary solution. Other advantages to AC/DC/AC converters is that they are stable in overload and no-load conditions, as well as they can be disengaged from a load without damage.[23]
Hybrid matrix converter: Hybrid matrix converters are relatively new for AC/AC converters. These converters combine the AC/DC/AC design with the matrix converter design. Multiple types of hybrid converters have been developed in this new category, an example being a converter that uses uni-directional switches and two converter stages without the dc-link; without the capacitors or inductors needed for a dc-link, the weight and size of the converter is reduced. Two sub-categories exist from the hybrid converters, named hybrid direct matrix converter (HDMC) and hybrid indirect matrix converter (HIMC). HDMC convert the voltage and current in one stage, while the HIMC utilizes separate stages, like the AC/DC/AC converter, but without the use of an intermediate storage element.[24][25]
Applications: Below is a list of common applications that each converter is used in.
Simulations of power electronic systems
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Output voltage of a full-wave rectifier with controlled thyristorsPower electronic circuits are simulated using computer simulation programs such as SIMBA, PLECS, PSIM, SPICE, MATLAB/simulink, and OpenModelica. Circuits are simulated before they are produced to test how the circuits respond under certain conditions. Also, creating a simulation is both cheaper and faster than creating a prototype to use for testing.[26]
Applications
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Applications of power electronics range in size from a switched mode power supply in an AC adapter, battery chargers, audio amplifiers, fluorescent lamp ballasts, through variable frequency drives and DC motor drives used to operate pumps, fans, and manufacturing machinery, up to gigawatt-scale high voltage direct current power transmission systems used to interconnect electrical grids.[27] Power electronic systems are found in virtually every electronic device. For example:
Motor drives are found in pumps, blowers, and mill drives for textile, paper, cement and other such facilities. Drives may be used for power conversion and for motion control.[28] For AC motors, applications include variable-frequency drives, motor soft starters and excitation systems.[29]
In hybrid electric vehicles (HEVs), power electronics are used in two formats: series hybrid and parallel hybrid. The difference between a series hybrid and a parallel hybrid is the relationship of the electric motor to the internal combustion engine (ICE). Devices used in electric vehicles consist mostly of dc/dc converters for battery charging and dc/ac converters to power the propulsion motor. Electric trains use power electronic devices to obtain power, as well as for vector control using pulse-width modulation (PWM) rectifiers. The trains obtain their power from power lines. Another new usage for power electronics is in elevator systems. These systems may use thyristors, inverters, permanent magnet motors, or various hybrid systems that incorporate PWM systems and standard motors.[30]
Inverters
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In general, inverters are utilized in applications requiring direct conversion of electrical energy from DC to AC or indirect conversion from AC to AC. DC to AC conversion is useful for many fields, including power conditioning, harmonic compensation, motor drives, renewable energy grid integration, and spacecraft solar power systems.
In power systems it is often desired to eliminate harmonic content found in line currents. VSIs can be used as active power filters to provide this compensation. Based on measured line currents and voltages, a control system determines reference current signals for each phase. This is fed back through an outer loop and subtracted from actual current signals to create current signals for an inner loop to the inverter. These signals then cause the inverter to generate output currents that compensate for the harmonic content. This configuration requires no real power consumption, as it is fully fed by the line; the DC link is simply a capacitor that is kept at a constant voltage by the control system.[15] In this configuration, output currents are in phase with line voltages to produce a unity power factor. Conversely, VAR compensation is possible in a similar configuration where output currents lead line voltages to improve the overall power factor.[16]
In facilities that require energy at all times, such as hospitals and airports, UPS systems are utilized. In a standby system, an inverter is brought online when the normally supplying grid is interrupted. Power is instantaneously drawn from onsite batteries and converted into usable AC voltage by the VSI, until grid power is restored, or until backup generators are brought online. In an online UPS system, a rectifier-DC-link-inverter is used to protect the load from transients and harmonic content. A battery in parallel with the DC-link is kept fully charged by the output in case the grid power is interrupted, while the output of the inverter is fed through a low pass filter to the load. High power quality and independence from disturbances is achieved.[15]
Various AC motor drives have been developed for speed, torque, and position control of AC motors. These drives can be categorized as low-performance or as high-performance, based on whether they are scalar-controlled or vector-controlled, respectively. In scalar-controlled drives, fundamental stator current, or voltage frequency and amplitude, are the only controllable quantities. Therefore, these drives are employed in applications where high quality control is not required, such as fans and compressors. On the other hand, vector-controlled drives allow for instantaneous current and voltage values to be controlled continuously. This high performance is necessary for applications such as elevators and electric cars.[15]
Inverters are also vital to many renewable energy applications. In photovoltaic purposes, the inverter, which is usually a PWM VSI, gets fed by the DC electrical energy output of a photovoltaic module or array. The inverter then converts this into an AC voltage to be interfaced with either a load or the utility grid. Inverters may also be employed in other renewable systems, such as wind turbines. In these applications, the turbine speed usually varies, causing changes in voltage frequency and sometimes in the magnitude. In this case, the generated voltage can be rectified and then inverted to stabilize frequency and magnitude.[15]
Smart grid
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A smart grid is a modernized electrical grid that uses information and communications technology to gather and act on information, such as information about the behaviors of suppliers and consumers, in an automated fashion to improve the efficiency, reliability, economics, and sustainability of the production and distribution of electricity.[31][32]
Electric power generated by wind turbines and hydroelectric turbines by using induction generators can cause variances in the frequency at which power is generated. Power electronic devices are utilized in these systems to convert the generated ac voltages into high-voltage direct current (HVDC). The HVDC power can be more easily converted into three phase power that is coherent with the power associated to the existing power grid. Through these devices, the power delivered by these systems is cleaner and has a higher associated power factor. Wind power systems optimum torque is obtained either through a gearbox or direct drive technologies that can reduce the size of the power electronics device.[33]
Electric power can be generated through photovoltaic cells by using power electronic devices. The produced power is usually then transformed by solar inverters. Inverters are divided into three different types: central, module-integrated, and string. Central converters can be connected either in parallel or in series on the DC side of the system. For photovoltaic "farms", a single central converter is used for the entire system. Module-integrated converters are connected in series on either the DC or AC side. Normally several modules are used within a photovoltaic system, since the system requires these converters on both DC and AC terminals. A string converter is used in a system that utilizes photovoltaic cells that are facing different directions. It is used to convert the power generated to each string, or line, in which the photovoltaic cells are interacting.[33]
Power electronics can be used to help utilities adapt to the rapid increase in distributed residential/commercial solar power generation. Germany and parts of Hawaii, California, and New Jersey require costly studies to be conducted before approving new solar installations. Relatively small-scale ground- or pole-mounted devices create the potential for a distributed control infrastructure to monitor and manage the flow of power. Traditional electromechanical systems, such as capacitor banks or voltage regulators at substations, can take minutes to adjust voltage and can be distant from the solar installations where the problems originate. If voltage on a neighborhood circuit goes too high, it can endanger utility crews and cause damage to both utility and customer equipment. Further, a grid fault causes photovoltaic generators to shut down immediately, spiking the demand for grid power. Smart grid-based regulators are more controllable than far more numerous consumer devices.[34]
In another approach, a group of 16 western utilities called the Western Electric Industry Leaders called for the mandatory use of "smart inverters." These devices convert DC to household AC and can also help with power quality. Such devices could eliminate the need for expensive utility equipment upgrades at a much lower total cost.[34]
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The work of solid-state relays and solid-state modules is inseparable from their internal power electronic devices, so it is necessary to introduce some basic knowledge of power electronic devices. Through this article, you will learn What are the power electronic devices? How do they work? What are their basic characteristics? How to use them?
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In the power equipment or power system, the main circuit is used to realize the change or control of electric energy, and the Power Electronic Device (PED) is the core of the main circuit. Earlier, power electronic devices included power vacuum devices (such as mercury arc rectifiers, thyratron) and power semiconductor devices (such as power diodes, thyristors). Due to the obvious advantages of power semiconductor devices in cost and performance, they have gradually replaced the position of power vacuum devices, thus so-called power electronic devices nowadays usually refer to power semiconductor devices with silicon as the main material.
Compared with information semiconductor devices (or Information Electronic Devices, IED) that also use silicon as the main material, power electronic devices have the following characteristics:
● Power electronic devices have a large power processing capacity, but due to the large power loss, they need to be equipped with a radiator for cooling.
● Power electronic devices often require information electronic devices to provide control signals.
● Power electronic devices generally work in the switching state, not in the amplifying state, in order to reduce their power consumption.
Due to the characteristics of semiconductors, power electronic devices will inevitably produce certain power losses during operation. These power losses will not only reduce the conversion efficiency of electrical energy, but also cause permanent damage to the power electronic devices due to overheating. The main losses of power electronic devices can be divided into on-state loss, off-state loss, and switching loss. The on-state loss is the loss caused by the on-state voltage drop, and if the switching frequency of the power electronic device is not high, the on-state loss will account for a high proportion of the total loss. The off-state loss is the loss caused by the off-state leakage current, and the off-state loss usually accounts for a very small proportion of the total loss and is often ignored. The switching loss refers to the loss generated during the switching process of power electronic devices, and the switching loss is greatly affected by the switching frequency -- the higher the switching frequency, the greater the proportion of the switching loss in the total loss.
Because of the differences in the materials and structures used, the performance of different types of power electronic devices may be completely different. Therefore, before choosing power electronic devices, you must first understand their categories and characteristics to give full play to their advantages.
According to the degree of control, power electronic devices can be divided into uncontrollable type, half-controlled type, and fully-controlled type.
According to the drive circuit signal, power electronic devices can be divided into current drive type, and voltage drive type. The working state of the current-driven devices is controlled by the input current, such as thyristor, GTO, GTR, and etc. The working state of the voltage-driven devices is controlled by the electric field effect generated by the input voltage, such as MOSFET, JFET, IGBT, and etc.
According to the carrier involved in the conduction process, power electronic devices can be divided into unipolar type, and bipolar type. When unipolar type devices work, only one type of carrier (free electron or hole) participates in the conduction process, such as MOSFET, JFET, SIT, and etc. When bipolar type devices work, free electrons and holes are involved together in the conduction process, such as thyristor, GTO, GTR, IGBT, SITH, TRIAC, RCT, LTT, and etc.
Most of the voltage-driven devices are unipolar type, and most of the current-driven devices are bipolar type. Voltage-driven devices usually have the characteristics of high input impedance, low driving power, simple driving circuit, and high operating frequency. Current-driven devices usually have a conductance modulation effect, so the on-state voltage drop and the conduction loss is small, but the operating frequency is low, the required driving power is large, and the driving circuit is more complicated.
Uncontrollable devices refer to devices that cannot be turned on and off by control signals, so no drive circuit is required. Uncontrollable devices generally refer to power diodes, and their basic structure and working principle are very similar to diodes. The power diode is encapsulated by a large-area PN junction and lead wires at both ends. According to the shape, power diodes can be divided into bolt type and flat type. According to the carriers involved in the conduction process, power diodes can be divided into unipolar power diodes and bipolar power diodes. Power diodes are widely used in power equipment in various fields because of their simple structure and low price. Especially fast recovery diodes and Schottky barrier diodes have an irreplaceable position in the rectification and inverter of low voltage, intermediate frequency and high frequency fields. With the development of modularization and integration technology, modular power diodes are becoming more and more common in the market (click to view more power diode modules).
The essence of the power diode is the PN junction formed by the contact between the P-type semiconductor and the N-type semiconductor. Therefore, in order to understand the working principle of the power diode, it is necessary to understand the basic characteristics and working principle of the PN junction.
The power diode is composed of a P-type semiconductor and an N-type semiconductor. N-type semiconductors and P-type semiconductors are composed of doped intrinsic semiconductors -- the concentration of free electrons in N-type semiconductors is high; the concentration of holes in P-type semiconductors is high. The connection area between the P-type semiconductor and the N-type semiconductor is called the PN junction. Free electrons and holes are also called free carriers (referred to as carriers) . The movement and recombination of free electrons in the semiconductor will inevitably lead to the generation and recombination of holes. From a macro point of view, this process is more like negatively charged free electrons and positively charged holes moving in opposite directions in the semiconductor at the same time. The movement of free electrons and holes in semiconductors is very fast and random, so it is almost impossible to predict the trajectory of a certain free electron or hole and accurately know its position at a certain moment. However, the movement of a large number of holes and free electrons is not without rules.
Majority carriers in semiconductors will diffuse from high-concentration regions to low-concentration regions -- the majority carrier free electrons (nn) in the N region diffuse from the high-concentration N region through the PN junction to the low-concentration P region; at the same time, the majority carrier holes (pp) in the P region diffuse from the high concentration P region through the PN junction to the low concentration N region. The carriers near the PN junction are depleted due to the diffusion movement, leaving only space charges (positive ions and negative ions) that cannot be moved, so this area is called the space charge region (also known as the depletion region). Since there are no free moving carriers in the space charge area, it is similar to an insulator. The space charge in the space charge region will generate a built-in electric field to prevent carriers from passing through the PN junction (the built-in electric field will be formed within a few nanoseconds after the PN junction is manufactured). Even so, there are still very few carriers that pass through the PN junction and become minority carriers in the opposite region -- free electrons become minority carriers in the P region (pn); holes become minority carriers in the N region (np). This phenomenon is called the quantum tunneling effect. The difference in carrier concentration on both sides of the space charge region produces a built-in potential difference (also called a built-in potential, or contact potential difference). Minority carriers will continue to diffuse into the lower concentration area. The average distance that the minority carriers can reach in the process of diffusion and recombination is called the diffusion length. The diffusion length is affected by the minority carrier lifetime -- the longer the minority carrier lifetime, the longer the diffusion length. When reaching the edge of the diffusion region, the minority carriers will pass through the PN junction and return to their original region under the action of the built-in electric field.
Single crystal semiconductor refers to a semiconductor with pure chemical composition, no impurities and no lattice defects, that is, intrinsic semiconductor, such as silicon (Si), germanium (Ge), gallium arsenide (GaAs), etc. The structure of single crystal semiconductors is very regular, and its macroscopic properties are anisotropic (in different directions, their physical properties are different). Single crystal semiconductors are the materials for most semiconductor devices.
Polycrystalline semiconductor refers to a semiconductor material composed of a large number of tiny single crystal semiconductor particles with different orientations. The structure of polycrystalline semiconductors does not have regularity, and their macroscopic properties are often isotropic (in different directions, their physical properties are the same). Polycrystalline semiconductors can be used to make narrow-film transistor switch matrices for solar cells, liquid crystal displays, and gate materials for MOSFET.
Energy band theory is a theory that uses quantum mechanics to study the movement of electrons inside a solid. In a coordinate system with energy as the ordinate, the energy of electrons in the crystal can be represented by a horizontal line -- the greater the energy, the higher the position of the line. Energy levels that are very close to each other within a certain energy range form the energy band. The vertical distance between the highest energy level and the lowest energy level in the energy band is called the energy band width. The position and width of the energy band are affected by the crystal type (such as metal, semiconductor, insulator), temperature and etc. The energy band of a semiconductor is shown in Figure 3.2a.
Full Band: It refers to the energy band that is completely occupied by electrons when T = 0K. The electrons in the full band are valence electrons (that is, electrons that are bound by the valence bonds on the crystal atoms and cannot move freely), so it does not have any conductivity.
Empty Band: It refers to the energy band that is not occupied by electrons when T = 0K. There are no electrons in the empty band, so it does not have any conductivity. The empty band becomes the conduction band when there are electrons in it.
Conduction Band: It refers to the energy band that is not completely occupied by electrons when T > 0K. The electrons in the conduction band are free electrons, so it has conductivity.
Valence Band: It refers to the energy band occupied by valence electrons when T > 0K. The electrons in the valence band are valence electrons, so it does not have any conductivity.
Forbidden Band: It refers to the energy range between the top of the valence band and the bottom of the conduction band. There is no energy level of shared electrons (that is, electrons shared by multiple atoms in the crystal) in the forbidden band, but there are energy levels of non-shared electrons (that is, localized electrons in impurities and defects). The forbidden band width (band gap) reflects the bondage degree of valence electrons or the strength of the valence bond, that is, the minimum average energy required for intrinsic excitation. The middle line of forbidden band is Fermi level.
The band gap of common metal materials is generally very small (the conduction band and the valence band even overlap each other), and their electrons can easily gain energy and transition to the conduction band at room temperature, so their conductivity is strong. The band gap of insulating materials is generally very large (usually greater than 9 electron volts, or 9 eV), and their electrons are difficultly to transition to the conduction band, so their conductivity is very poor. The band gap of semiconductor materials is between conductors and insulators (about 1-3 eV, for example, at room temperature, the band gap of germanium is 0.67 eV, and the band gap of silicon is 1.12 eV), so just give appropriate energy excitation (breaking the valence bond to make the valence electron transition to the conduction band to generate free electrons and holes) or change the band gap (reducing the energy required for electronic transitions) can make the semiconductor conductive.
Intrinsic excitation means that by giving certain excitation conditions, the electrons in the intrinsic semiconductor will cross the forbidden band from the lower energy band (full band or valence band) into the higher energy band (empty band or conduction band) and becomes the free electrons. It should be noted that the free electrons in the conduction band of the intrinsic semiconductor refer to the approximately free electrons in the solid, which can move freely in the entire solid, but cannot run out of the solid. A positively charged vacancy is formed due to the lack of an electron in the lower energy band, which is called the hole. Free electrons in the conduction band and holes in the valence band are collectively called electron-hole pairs. In intrinsic semiconductors, free electrons and holes generated by intrinsic excitation can move freely, so they are called free carriers, and their concentrations are equal to each other, and as the temperature rises, their concentration will increase exponentially. The directional movement of free electrons and holes will form electrons flow and holes flow. The free electrons in the conduction band will fall into the holes, causing the electron-hole pairs to disappear. This process is called recombination. The energy generated during recombination is released in the form of electromagnetic radiation (emitting photon) or thermal vibration of the lattice (emitting phonon). At a certain temperature, the generation and recombination of electron-hole pairs exist simultaneously and reach a dynamic equilibrium. At this time, the intrinsic semiconductor has a certain carrier concentration and thus it has a certain electrical conductivity. Through intrinsic excitation, more electron-hole pairs are generated, thereby increasing the carrier concentration, which can effectively increase the conductivity of the semiconductor. According to this principle, semiconductor devices such as semiconductor thermistors and semiconductor photo-resistors can be manufactured. The conductivity of intrinsic semiconductors at room temperature is small, and the carrier concentration is sensitive to temperature changes, so it is difficult to effectively control the semiconductor characteristics of their semiconductors through temperature.
Intrinsic excitation methods can generally be divided into intrinsic thermal excitation, intrinsic light excitation, and impact ionization intrinsic excitation.
Intrinsic Thermal Excitation: It refers to the infrared photons radiated by thermal motion of molecules as the temperature increases, which makes valence electrons to gain enough energy to break free from the bondage of valence bonds and become free electrons. The energy required for intrinsic thermal excitation is equal to the band gap. The thermal excitation efficiency at room temperature is usually very limited, because a very high temperature is required to allow enough carriers to transition to the conduction band.
Intrinsic Light Excitation: It refers to the photons radiated by light (generally referring to visible light or ultraviolet light), which makes valence electrons to gain enough energy to break free from the bondage of valence bonds and become free electrons. The energy required for intrinsic light excitation is greater than intrinsic thermal excitation. Since the photons of visible light have higher energy than the infrared photons usually generated by thermal motion, the energy of electrons after intrinsic light excitation is usually located at a higher position in the conduction band. Because the momentum of the photon can be ignored, the intrinsic light excitation does not change the momentum of the electron, so this process is also called the vertical transition. But if there is a phonon (referring to the simple harmonic vibration of the crystal lattice) involved, the momentum of the electron will change, so it is also called a non-vertical transition.
Impact Ionization Intrinsic Excitation: It refers to the collision and ionization of valence electrons by high-energy electrons (that is, free electrons accelerated by an electric field) to become free electrons. The average energy required for impact ionization intrinsic excitation is about 1.5 times the band gap. The electrons produced by the intrinsic excitation of impact ionization are ionized electrons, which are the truly free electrons that can leave the solid, and their energy exceeds the free electrons with the highest energy level in the conduction band.
The Fermi level is the energy level that has a 50% chance of being occupied by electrons at any temperature -- below the Fermi level, the farther the distance is, the greater the possibility of being occupied by the electrons; above the Fermi level, the farther the distance is, the less the possibility of being occupied by the electrons. For semiconductors, especially intrinsic semiconductors, the Fermi level is in the middle line of the forbidden band. When the temperature T = 0K, the full band is filled with electrons (the electron occupancy probability is 1), and the empty band has no electrons at all (the electron occupancy probability is 0), then their Fermi level is exactly at the middle line of the forbidden band (the electron occupancy probability is 1/2). Even when the temperature rises T> 0K, intrinsic excitation will produce electron-hole pairs, but since the number of electrons increased in the conduction band is equal to the number of electrons decreased in the valence band, the Fermi level is still in the middle line of the forbidden band (the electron occupancy probability is 1/2). Therefore, the position of the Fermi level of the intrinsic semiconductor does not change with temperature and is always at the middle line of the forbidden band. The Fermi-Dirac distribution of electrons can be calculated from the Fermi level and temperature, as shown in Figure 3.2b.
Generally, intrinsic semiconductor will be doped to introduce new energy levels to increase its conductivity. The doped semiconductor is more susceptible to external influences (such as light, temperature rise, etc.).
By doping the silicon crystal (or germanium crystal) with phosphorus element (or antimony element), the phosphorus atom (or antimony atom) will occupy the position of the silicon atom. Then a set of full energy levels will be added to the position in the forbidden band which is very close to the conduction band. The electrons on these energy levels can easily transition to the conduction band to become free electrons. Therefore, the phosphorus element (or antimony element) is called a donor impurity (or N-type impurity), and a semiconductor doped with an N-type impurity is called an N-type semiconductor.
By doping the silicon crystal (or germanium crystal) with boron element (or indium element), the boron atom (or indium atom) will occupy the position of the silicon atom. Then a set of empty energy levels will be added to the position in the forbidden band which is very close to the valence band. The electrons in the valence band can easily transition to these energy levels and leave holes in the valence band. Therefore, boron element (or indium element) is called acceptor impurities (or P-type impurities), and semiconductors doped with P-type impurities are called P-type semiconductors.
The essence of the working principle of the power diode is the unidirectional conductivity of the PN junction. In the case of constant external conditions (temperature, radiation, etc.), the external circuit will supplement the carriers consumed during the operation of the power diode, so the conductivity of the power diode is mainly affected by its internal carrier concentration.
Forward Conduction State: If a forward bias voltage is applied across the power diode, the majority carriers will move closer to the PN junction, which will narrow the space charge region and weaken the built-in electric field, but the PN junction will still maintain dynamic equilibrium state. Only when the forward bias voltage is greater than the built-in electric field, the dynamic equilibrium can be broken, and a superimposed electric field in the opposite direction of the drift current is generated. The diffusion current ID is greater than the drift current Id. The current flowing through the PN junction is the forward current IF. Due to the built-in potential difference, when the power diode is in the forward conducting state, an on-state voltage drop will be generated at both ends of it, which makes the power diode present a low impedance state. The on-state voltage drop of the power diode is not a fixed value, it is proportional to the current flowing.
Reverse Cut-off State: If a reverse bias voltage is applied across the power diode, the majority carriers will move away from the PN junction, which will widen the space charge region and enhance the built-in electric field. The dynamic equilibrium of the PN junction is broken, and a superimposed electric field in the same direction as the drift current is generated. The drift current Id is greater than the diffusion current ID. The current flowing through the PN junction is the reverse saturation current Isat. Because the number of minority carriers is too small, the reverse saturation current of the power diode is usually negligible, which makes the power diode present a high impedance state.
Reverse Breakdown State: If the reverse bias voltage across the power diode continues to increase to a certain critical value, the number of carriers in the power diode increases rapidly, causing the reverse current IR to increase significantly. In the reverse breakdown state, the power diode presents a no-impedance state, and its reverse current IR and reverse voltage UR are both very large. The reverse breakdown of PN junction is mainly divided into avalanche breakdown and Zener breakdown. Both of these breakdowns will increase the temperature of the PN junction, and eventually lead to thermal breakdown, which will cause permanent damage to the PN junction. If the cooling measures are done well enough, even if the power diode is reverse broken down, but the PN junction is not destroyed, then after limiting or closing the reverse voltage, the PN junction can still be restored to its original state.
Diffusion motion refers to the movement of majority carriers from a high-concentration area to a low-concentration area. The diffusion motion is determined by the concentration gradient. Drift motion refers to the movement of minority carriers returning to the original area under the action of the built-in electric field. The drift motion is determined by the built-in electric field. Diffusion motion and drift motion are the basic motions of carriers in the PN junction. The current generated by the diffusion motion is called the diffusion current ID, and the current generated by the drift motion is called the drift current Id. When there is no applied voltage (or the forward bias voltage is less than the built-in electric field), the diffusion motion will cause the built-in electric field to enhance and the drift current will increase; the drift motion will cause the built-in electric field to weaken and the diffusion current to increase. Finally, the diffusion current is equal to the drift current, and the PN junction will be in a dynamic equilibrium state where the total current is zero.
Avalanche breakdown usually occurs in a low-doped PN junction with a wide depletion layer. Due to the wide depletion layer, when the reverse bias is large, the free electrons in the semiconductor will continuously accelerate under the electric field force, and obtain a large number of kinetic energies. These high-energy electrons collide with valence electrons, freeing them from the bondage of valence bonds and generating new electron-hole pairs. These newly generated free electrons continue to repeat this process under the action of the electric field force, causing the free carriers in the semiconductor to increase rapidly like an avalanche, leading to a sharp increase in drift current. The essence of the avalanche breakdown is the impact ionization excitation, so the avalanche breakdown voltage is usually high (generally higher than 6V). The avalanche breakdown voltage increases with the increase of temperature, mainly because the irregular thermal movement of carriers increases with the increase of temperature, so a larger reverse voltage is required to provide a large enough electric field to make the carriers do directional acceleration motion.
Zener breakdown usually occurs in a highly doped PN junction with a narrow depletion layer. Due to the narrow depletion layer, there is not enough space for free electrons to accelerate, so avalanche breakdown will not occur. At the same time, due to the narrow depletion layer, even if the reverse bias is not large, a strong electric field can still be generated in the PN junction, pulling electrons out of the valence bond, and generating new electron-hole pairs. This phenomenon is also called field-induced excitation. Field-induced excitation will greatly increase the number of carriers in the semiconductor, thereby significantly increasing the drift current. Zener breakdown voltage is usually low (generally less than 4V). The Zener breakdown voltage decreases with increasing temperature, mainly because the electrons in the valence bond become more active with increasing temperature, so it is more easily for the electric field force to pull them out.
The amount of charge in the PN junction changes with the applied voltage, exhibiting a capacitance effect. This capacitance is called the junction capacitance CJ (also known as differential capacitance). According to the different generation mechanism and function, the junction capacitance CJ can be divided into the barrier capacitance CB and the diffusion capacitance CD, and they conform to the calculation formula CJ=CB+CD. Both the barrier capacitance and the diffusion capacitance are non-linear capacitance.
The narrow layer of ions in the space charge region (depletion layer) forms the barrier region. The number of space charges in the barrier area changes with the applied bias voltage, which is equivalent to the charge and discharge effect of the capacitor -- when the forward bias voltage increases, the barrier area decreases, which is equivalent to storing free electrons or holes into the barrier area; when the forward bias decreases, the barrier area increases, which is equivalent to taking out free electrons or holes from the barrier area. The equivalent capacitance of the barrier region is called the barrier capacitance CB. If the frequency of the applied bias voltage is higher, the effect of the barrier capacitance is more obvious. Regardless of low-frequency operation or high-frequency operation, the barrier capacitance may deteriorate the unidirectional conductivity of the semiconductor device, or even fail to work. In fact, the maximum operating frequency of a semiconductor device is often determined by the barrier capacitance. It is worth noting that the barrier capacitance is the capacitance effect related to the majority carrier (pp and nn), and neither the forward bias nor the reverse bias can be ignored. In forward bias, when the forward voltage is low, the barrier capacitance is much larger than the diffusion capacitance, so the barrier capacitance is the main component of the junction capacitance, CJ ≈ CB.
When the PN junction is forward biased, the built-in electric field is weakened, and the drift motion of minority carriers is weakened. The diffusion current is greater than the drift current. Therefore, the carriers that diffuse to the opposite area will accumulate at the barrier boundary to form a certain concentration of non-equilibrium minority carriers (pn and np) -- the closer to the PN junction, the higher the concentration; the farther away from the PN junction, the lower the concentration. The amount of charge of such non-equilibrium minority carriers changes with the forward bias, which is equivalent to the charge and discharge effect of the capacitor -- when the forward bias increases, the non-equilibrium minority carriers are increased, which is equivalent to the charging of the capacitor; when the forward bias decreases, the non-equilibrium minority carriers are reduced, which is equivalent to the discharging of the capacitor. The equivalent capacitance at the boundary of the barrier region is called the diffusion capacitance CD. The diffusion capacitance has a great influence on the switching speed of the PN junction when working at low frequencies, and can be ignored when working at high frequencies. In forward bias, if the forward voltage is high, the diffusion capacitance is much larger than the barrier capacitance, so the diffusion capacitance is the main component of the junction capacitance, CJ ≈ CD. In reverse bias, there are too few non-equilibrium minority carriers, and the diffusion capacitance can be ignored, so the barrier capacitance is the main component of the junction capacitance, CJ ≈ CB.
The maximum forward average current IF(AV) is the rated current of the power diode, which refers to the average value of the maximum power frequency half-wave sine current allowed to flow through the power diode under the specified case temperature TC and heat dissipation conditions. If it exceeds IF(AV), the diode will be burned out. Since the waveform of some power diodes is not necessarily a half-sine wave, and some power diodes do not have resistance characteristics, IF(AV) is defined according to the thermal effect of current, that is, find a resistor with similar heat generation according to the principle of equal effective value. Considering that the heat dissipation conditions will affect the ability of the power diode to withstand current, it is recommended to leave a certain margin to avoid damage to the power diode due to heat dissipation problems.
The threshold voltage UTO (also known as the dead zone voltage) is the lowest forward voltage at which the power diode can be turned on. The threshold voltage is the lowest forward voltage drop of the power diode. The threshold voltage of germanium crystal is about 0.1V; the threshold voltage of silicon crystal is about 0.5V.
The on-state voltage drop UCO (also known as the conduction voltage) is the forward voltage drop when the power diode is turned on and works stably. Ideally, the on-state voltage drop of the power diode is equal to the built-in potential. The built-in potential is related to the degree of semiconductor doping and is approximately equal to the half of the band gap. The on-state voltage drop of the power diode is proportional to the current flowing. The on-state voltage drop of germanium crystals is usually around 0.1-0.3V; the on-state voltage drop of silicon crystals is usually around 0.5-0.8V.
The maximum forward voltage drop UFM is the forward voltage drop corresponding to the maximum forward average current IFM(AV) at a specified temperature.
When an appropriate reverse voltage is applied, a very small leakage current will be generated, which is called the reverse saturation current Isat. The reverse saturation current is generated by the drift motion of minority carriers, so it is greatly affected by temperature.
The reverse repetitive peak voltage URRM (also known as the maximum reverse voltage URM) is the rated voltage of the power diode, which refers to the highest reverse voltage that the power diode can withstand repeatedly applied. If it exceeds this value, the power diode will be reversed and damaged. Taking into account the overvoltage in the circuit and other factors, when using power diodes, there should usually be a double margin. For example, a power diode with a rated voltage of 1000V can only be used as a 500V power diode.
The reverse recovery process is caused by the capacitance effect of the power diode. When the switch transitions from the on state to the off state, the power diode needs to release the charge stored in the junction capacitance before blocking the reverse current. This discharge time is called the reverse recovery time trr, that is, the time from when the forward conduction current is zero to when it enters the fully turn-off state. The reverse recovery time of power diodes of different specifications is different, so you need to fully consider when designing the circuit, otherwise it may cause unnecessary trouble. For example, the reverse recovery time of a power diode is Trr. If a continuous PWM wave with a period of T1 (T1<Trr) passes through the power diode, the PWM wave cannot be blocked when the power diode is reversely biasing.
The junction temperature TJ refers to the average temperature of the PN junction. The maximum operating junction temperature TJM refers to the highest average temperature that the PN junction can withstand without damage (usually the highest junction temperature of germanium transistors is about 75°C, and the highest junction temperature of silicon transistors is about 150°C). Temperature has a very significant impact on the working characteristics of power diodes, so sufficient heat dissipation conditions must be provided to avoid damage to the power diode due to overheating.
The maximum operating frequency fM is the upper turn-off frequency of the diode. If the frequency is too high, the power diode will easily lose its ability to block reverse current due to the capacitive effect. At the same time, if the frequency is too high, it will also cause the power diode to be burned due to the increase of on-state power consumption.
The surge current IFSM refers to the maximum continuous overcurrent of one or several power-frequency cycles that the power diode can withstand.
The static characteristic of the power diode mainly refers to the volt-ampere characteristic curve of the power diode, shown in Figure 6.
When a forward bias voltage is applied to both ends of the power diode, the power diode will not be turned on immediately. Only when the forward voltage is greater than the threshold voltage UTO of the power diode, the power diode will be turned on. At this time, the forward current IF begins to increase significantly, until the power diode is in a stable conduction state, at which time the diode's conduction voltage is UCO. If the forward current reaches IFM, the corresponding voltage drop is UFM, and the power diode will be burned out due to excessive current.
When a reverse bias voltage is applied to both ends of the power diode, the power diode will not conduct, but will generate a small constant value current, that is, reverse leakage current. When the reverse voltage reaches the reverse UBR of the power diode, the power diode will be reversely broken down, and the reverse current will become very large at this time.
The dynamic characteristics of the power diode refer to its switching characteristics, that is, the voltage-current characteristic of the power diode during the transition between on-state and off-state. Because of the junction capacitance, the voltage-current characteristics of power diodes change with time.
The dynamic characteristic of the power diode in the turn-on process is shown in Figure 7. When the voltage changes from zero bias to forward bias, the forward current IF of the power diode will increase from 0 to IF1. Due to the large di/dt, under the action of the line inductance, a forward peak voltage UFP will be generated at both ends of the power diode. After a certain period of time, the forward voltage UF will gradually drop from UFP to the stable voltage UF1 (that is, the on-state voltage drop). In this process, the time when the forward current rises from 0 to IF1 is called the forward recovery time tfr.
The dynamic characteristic of the power diode in the turn-off process is shown in Figure 8. Due to the junction capacitance, even if the forward bias is converted to the reverse bias, the power diode will not be turned off immediately, but it will take a period of time to regain the reverse blocking capability.
When the power diode is switched from forward bias to reverse bias at tF, and the forward current IF decreases rapidly, and drops to 0 at t0, and diF/dt is large. From t0 to t1, the current not only does not disappear, but becomes the reverse current IR and increases rapidly until it reaches the maximum value IRP. This time period is called the delay time td. From t1 to t2, the reverse current begins to drop sharply to a very small value. This time period is called the fall time tf. From t2, the reverse current begins to slowly decrease until it drops to 0 (in fact, there is still a very small reverse leakage current). The time from t0 to t2 is called the reverse recovery time trr, during which the power diode is reverse conducting. The reverse recovery time trr determines the operating frequency of the power diode. If the operating frequency of the external circuit is too high, the power diode cannot enter the reverse cut-off state when reverse biased, and there is a large reverse current, which is equivalent to the power diode losing its reverse blocking ability.
Before the reverse current rises to the maximum value, the voltage across the power diode drops rapidly from the on-state voltage drop UF1 to 0. At the same time, since tf is usually very short, diR/dt is very large. Under the action of the line inductance, a reverse peak voltage URP is quickly generated at both ends of the power diode, and then it begins to drop to a stable value UR1. The reverse peak voltage is usually very large and may break down the power diode. Therefore, increasing the proportion of tf in trr will help reduce the reverse peak voltage. The recovery coefficient (Sr=tf/td) is usually used to express the softness of the reverse recovery characteristics of the power diode.
General purpose diodes (GPD), also known as rectifier diodes, have a long recovery time, high forward current rating and reverse voltage rating. They are mostly used in rectifier circuits with low switching frequency (below 1kHz), and generally cannot be used in medium and high frequency circuits.
The internal structure of the fast recovery diode (FRD) is different from that of the general purpose diode. It adds a base area I between the P-type and N-type silicon materials to form a P-I-N structure. Because the base area is very thin and the reverse recovery charge is small, it not only greatly reduces trr and the transient forward voltage drop, but also improves its reverse voltage withstand capability. The recovery time of the fast recovery diode is very short (trr>100ns, usually several hundred ns), its forward voltage drop is about 0.6V, the forward current is several amperes to several thousand amperes, and the reverse peak voltage can reach several hundred to several thousand volts. Ultra-fast recovery diodes, also known as fast recovery epitaxial diodes (FRED), have a further reduced reverse recovery charge, so they have a shorter recovery time (trr<100ns, as low as 20~30ns). The forward voltage drop of ultra-fast recovery diodes is also very low (about 0.9V), but its reverse withstand voltage capability is usually not high (less than 1200V).
Schottky Barrier Diode (SBD) is a kind of power diode based on the barrier formed by the contact between metal and semiconductor. Compared with general purpose diodes and fast recovery diodes, Schottky barrier diodes have the advantages of short reverse recovery time, no obvious forward voltage overshoot, and high reverse withstand voltage, but their reverse leakage current is large. The forward voltage drop of a Schottky barrier diodes is affected by the reverse withstand voltage -- if the reverse withstand voltage increases, the forward voltage drop will increase significantly. But when the reverse withstand voltage is low, the forward voltage drop of Schottky barrier diodes is significantly lower than that of general purpose diodes and fast recovery diodes, so the switching loss and on-state loss are very low. Therefore, Schottky barrier diodes are usually used in rectifier circuits below 200V. However, it should be noted that Schottky barrier diodes are very sensitive to temperature, so their operating temperature must be strictly limited.
Before introducing half-controlled devices and full-controlled devices, it is necessary to briefly introduce bipolar junction transistors (BJT).
The transistor (also known as semiconductor transistor or Bipolar Junction Transistor, BJT) is a bipolar device with three terminals and two PN junctions. The transistor is one of the basic components of semiconductor devices and also one of the core components. Since its birth in the 1940s, the transistor has completely changed the structure of electronic circuits, triggered a solid-state revolution, and promoted the emergence of integrated circuits and large-scale integrated circuits. The transistor has a current amplifying function, and can control a large change in collector current with a very small change in base current, so it is often used as a contact-less switch in electronic circuits. The switching frequency of the transistor is high, and there is no mechanical service life, so it has a significant advantage over electromagnetic relays and mechanical switches.
The transistor is a three-layer semiconductor structure, which has one more PN junction than Power Diode. These two closely spaced PN junctions divide the transistor into three parts with different areas and doping concentrations -- the base region is very thin (3-30μm) and the doping concentration is low; the area of emitter region is small and the doping concentration is high; the area of collector region is large and the doping concentration is low. The PN junction between the collector region and the base region is called the collector junction J1. The PN junction between the emitter region and the base region is called the emitter junction J2.
According to the material, transistors can be divided into silicon transistors and germanium transistors. According to the doping composition, transistors can be divided into PNP transistors and NPN transistors -- under forward bias, the emitter region of PNP transistors emits holes, and its direction is the same as the direction of current, so the arrow in the electrical symbol goes from the emitter to the base; under forward bias, the emitter of the NPN transistors emits free electrons, and its direction is opposite to the direction of the current, so the arrow in the electrical symbol goes from the base to the emitter.
Take the NPN transistor as an example. The NPN transistor can be regarded as two equivalent diodes (VD1 and VD2), as shown in Figure 11, a. Because the N- region of VD1 has low a doping concentration and a large area, it is not prone to avalanche breakdown, so it can withstand a large reverse voltage. But in forward bias, the forward current of VD1 is very small, so VD1 is very suitable for working in reverse cut-off state. VD1 will produce a reverse saturation current ICBO when VD1 works in the reverse state, but the doping concentration of N- region and P region is very low, so the ICBO is very small. Because the N+ region of VD2 has a high doping concentration and a small area, it is prone to avalanche breakdown, so its reverse withstand voltage capability is very poor. But in forward bias, VD1 can generate a very large forward current, so VD2 is very suitable for working in the forward conduction state. When VD2 works in the forward state, it will generate two currents, one is the current IEP generated by the holes flow of the P region, and the other is the current IEN generated by the electrons flow of the N+ region. Since the doping concentration of the P region of VD1 is lower than that of the N+ region, IEN is greater than IEP. When we know the working principle of these two equivalent diodes, it is easy to understand the working principle of the NPN transistor.
Connect the NPN transistor through the common emitter connection method -- a collector power supply EC and collector resistance RC are connected in series to the collector and the emitter; a base power supply EB and a base resistance RB are connected in series to the base and the emitter. In this circuit, current flows into the NPN transistor from the collector and base, and flows out of the NPN transistor from the emitter -- the total current flowing in from the collector is the collector current IC; the total current flowing in from the base is the base current IB; the total current flowing out from the emitter is the emitter current IE. The linear relationship between IC and IE is common base current gain α, and the linear relationship between IC and IB is common emitter current gain β. It is worth noting that due to the difference in doping concentration, the collector junction J1 is not suitable for forward bias, and the emitter junction J2 is not suitable for reverse bias. If the collector and emitter are reversely connected, the possibility of NPN transistor breakdown will increase significantly.
Cut-off State: The structure of NPN makes there is always a PN junction in a reverse bias state. When no voltage is applied to the base, even if a large voltage (less than breakdown voltage BVCEO) is applied to the collector and the emitter, the NPN transistor cannot be turned on (but there is a small leakage current ICEO).
Active State: VD1 and VD2 must work at the same time to turn on the NPN transistor, so a certain voltage needs to be applied to the base to make J1 reverse biased (UBC<0) and J2 forward biased (UBE> UTO). When the NPN transistor is turned on, its internal current is a bit different from when the equivalent diodes worked separately, as shown in Figure 11, b. The free electrons injected from the N+ region into the P region do not completely recombine with the holes in the P region. Due to the reverse bias of J2, a part of the free electrons will pass through the P region and be directly injected into the N- region, and generate a reverse current ICN. When the NPN transistor is working in an active state, a small change in the base current IB of will cause a large change in the collector current IC. This phenomenon is called conductance modulation effect. This phenomenon is like a tiny input current being amplified into a huge output current, so the active state is also known as the amplification state.
Saturation State: With the increase of IB, the concentration of holes in the P region decreases, IEP decreases, and the depletion region of J1 keeps increasing. At the same time, since the free electrons injected from the N+ region into the P region are getting less and less, IBN has dropped to near the minimum value, and the amplification effect of IB on IC has begun to weaken. When IB and IC no longer have a linear relationship, the NPN transistor begins to enter a saturated state. At this time, as IB increases, IC slowly increases, and the saturation depth of the NPN transistor also begins to deepen. When almost all the free electrons in the N+ region are injected into the N- region, the base potential is the same as the collector potential (UBC=0). At this time, the NPN transistor is in a deep saturation state, and IC is completely unaffected by IB. It should be noted that as the depletion region of J1 increases, the possibility of avalanche breakdown on J1 also increases.
IC = ICN + ICBO, (1)
IB = IBN + IEP - ICBO, (2)
IE = IC + IB = ICN + IBN + IEP, (3)
because IC > 0, then we get IE / IC = IB / IC + 1; (4)
α = ICN / IE = (IC - ICBO) / IE, (5)
β = ICN / (IB + ICBO) = (IC - ICBO) / (IB + ICBO), (6)
because IC > IB >> ICEO >> ICBO ≈ 0, if ignoring all the leakage current, we can get α ≈ IC / IE, β ≈ IC / IB, (7)
then we can get 1/α = 1/β + 1, (8)
so the relation between α and β is: α = β / (1 + β), β = α / (1 - α). (9)
Both the collector junction reverse saturation current ICBO and the penetration current ICEO are unavoidable leakage currents in the transistor. By opening the emitter of the transistor (IE=0) and applying voltage to the collector and the base, the value of ICBO can be measured, as shown in Figure 12, a. By opening the base of the transistor (IB=0) and applying voltage to the collector and the emitter, the value of ICEO can be measured, as shown in Figure 12, b.
The generation mechanism of the collector junction reverse saturation current ICBO is shown in Figure 11, a.
The generation mechanism of the penetration current ICEO is as follows: Under the action of an external electric field, the majority carriers in the collector region move away from the PN junction, widening the space charge region, and the built-in electric field of the collector junction J1 is enhanced, which is conducive to drift motion; the majority carriers in the emitter region move closer to the PN junction, narrowing the space charge region, and the built-in electric field of the emitter junction J2 is weakened, which is not conducive to drift motion. Therefore, under the action of the built-in electric field, the minority carriers in the base region drift to the collector region through J1. At the same time, the minority carriers in the collector region drift to the base region through J1, part of which participates in the recombination of the base region, and the other part diffuses to the emitter region through J2. Due to the low doping concentration of the base region, the proportion of minority carriers participating in the recombination of the base region is very low. It is not difficult to find that this process is very similar to the generation mechanism of IEN when the transistor is turned on. Therefore, there is a linear relationship between ICEO and ICBO, ICEO = (1 + β) * ICBO. However, due to the low doping concentration of the collector region and the base region, the value of ICEO is very low and can usually be ignored. The ICEO of silicon transistors is generally less than 100nA; the ICEO of germanium transistors is generally less than 100μA.
Conductance (G) is the reciprocal of resistance, and the unit is Siemens (S). Conductance modulation effect (also known as the base region conductivity modulation effect, or Webster effect) is one of the basic characteristics of bipolar transistors (BPT), which refers to the phenomenon that the conductivity of the base region increases significantly (or the resistivity of the base region decreases significantly) when the working current of the bipolar transistor is large. Except BJT, other bipolar transistors such as SCR, GTO, GTR and parasitic transistors in IGBT all have conductivity modulation effect. In addition to the Webster effect, when the working current of the bipolar transistor is large, the Early effect (the phenomenon that the changes of the collector junction voltage will lead to the changes of the width of the base region) and the Kirk effect (the phenomenon that the width of the base region increases) will also appear.
Common base current gain α (the full name is "hybrid parameter forward current gain, common base", HFB), which is determined by the emitter efficiency factor and the base region transport factor, α = FE * FB. When the base is zero-biased (UBC = 0), the base short-circuit amplification factor α0 is determined by the emitter efficiency factor, the base region transport factor, the collector efficiency factor and the avalanche multiplication factor, α0 = FE * FB * FC * M.
The emitter efficiency factor FE is the ratio of the electron current IEN injected into the base region to the emitter current IE, FE = IEN / IE = IEN / (IEN + IEP) = 1 / [1 + (IEP / IEN) ]. By reducing the doping concentration of the base region, the total amount of impurities in the base region is much smaller than the total amount of impurities in the emitter region, which can effectively increase the number of minority carriers injected into the base region from the emitter region. The closer the ratio of IEP to IEN is to 0, the higher the emission efficiency of the transistor.
The base region transport factor FB is the ratio of the electron current ICN that reaches the collector region to the electron current IEN injected into the base region, FB = ICN / IEN. By reducing the width of the base region, the time that carriers from the emitter region stay in the base region can be effectively shortened, thereby increasing the number of minority carriers that transit the base region. The smaller the width of the base region, the smaller the recombination loss of electrons from the emitter region in the base region.
The collector efficiency factor FC is the ratio of the collector current IC to the electron current ICN that reaches the collector region, FC=IC/ICN.
The avalanche multiplication factor M is used to describe the avalanche multiplication effect when the reverse voltage of the collector junction increases to close to the avalanche breakdown voltage. It is usually estimated with the following formula, M = 1 / [1 - (V / VB) ^n], n is determined by the material of the PN junction (silicon: n=1.5-4; germanium: n=2.5- 8); VB is the reverse breakdown voltage of the collector J1; V is the voltage across the collector junction. When the absolute value of V tends to the absolute value of VB, M tends to infinity, and avalanche breakdown will occur in the PN junction.
Generally, hFB(α) is used to express the common base DC current gain, hFB(α) = IC / IE, and its range is usually 0.95-0.99; hfb(α) is used to express the common base AC current gain, hfb(α) = ΔIC / ΔIE. In general, hfb(α) ≈ hFB(α).
Common emitter current gain β (the full name is "hybrid parameter forward current gain, common emitter", HFE) is the ratio of collector current to base current, and its value is usually much larger than 1. Generally, hFE(β) is used to express the common emitter DC current gain, hFE(β) = IC / IB, which can be measured directly by a multimeter; hfe(β) is used to express the common emitter AC current gain, hfe(β) = ΔIC / ΔIB. The current amplification factor (or forward current gain) of the transistor usually refers to the common emitter current gain β.
Common collector current gain γ (the full name is "hybrid parameter forward current gain, common collector", HFC) is the ratio of emitter current to base current. Generally, hFE(γ) is used to express the common collector DC current gain, hFC(γ) = IE / IB; hfc(γ) is used to express the common collector AC current gain, hfc(γ) = ΔIE / ΔIB. This parameter is rarely used in normal times.
The threshold voltage UTO is the voltage that triggers the conduction of the emitter junction of the transistor.
The characteristic frequency fT is also called the gain bandwidth product, which can be defined as the operating frequency of the transistor when β=1. If the operating frequency f0 and the high-frequency current amplification factor β are known, the characteristic frequency fT can be obtained, fT=β* f0. As the operating frequency increases, the magnification will decrease. If the operating frequency of the transistor is equal to the characteristic frequency (f0 = fT), the transistor completely loses the current amplification function; if the operating frequency of the transistor is greater than the characteristic frequency (f0> fT), the transistor will not work normally.
The maximum operating voltage UCEM is the rated voltage of the transistor. When the maximum operating voltage UCEM is exceeded, the transistor will be broken down.
The maximum collector allowable current ICM is the rated current of the transistor. It is usually specified that the collector current IC corresponding to when the current gain β drops by half from the maximum value is ICM. In order to ensure the safety of use, it is generally necessary to leave a double margin.
The maximum collector dissipation power PCM is the power at which the transistor reaches the highest junction temperature under the highest operating temperature (usually 25°C). When the transistor reaches the maximum junction temperature, its internal PN junction structure will be permanently destroyed.
The relationship between the parameters of the transistor during stable operation (these parameters are usually fixed values or changing slowly) is called static characteristics. The relationship between the parameters of the transistor during the turn-on process and the turn-off process (these parameters are usually changing sharply) is called dynamic characteristics. If there is only a DC signal in the input signal of the transistor, it is called DC operation (or static operation). If there is an AC signal in the input signal of the transistor, it is called AC operation (or dynamic operation).
For the NPN transistor (common emitter connection), its input is the base and the output is the collector, so its input current is IB, the input voltage is UBE, and the output current is IC (output from the resistance RC in the output circuit), the output voltage is UCE.
The static characteristics of the transistor are divided into input characteristics (relationship between input current and input voltage), output characteristics (relationship between output current and output voltage), temperature (the influence of temperature on input characteristics and output characteristics) and safe operating area (stable operating conditions of the transistor).
The input characteristic of the transistor is similar to the forward input characteristic of the power diode, as shown in Figure 13.
When UCE is fixed value and UBE>UTO, the base current IB increases with the increase of UBE.
When UCE increases, UTO increases and the input characteristics curve moves to the right. This is because with the increase of UCE, part of the carriers that should be injected into the base region from the emitter region pass through the base region and are directly injected to the collector region, so the carrier concentration in the base region is too low to open the emitter junction (that is, the diffusion current of the emitter junction is less than or equal to the drift current). Therefore, it is necessary to increase UBE to make more carriers be injected into the base region from the emitter region (that is, the threshold voltage increases), and the input characteristic curve also moves to the right. When UCE increases to a certain extent, most of the carriers that can be injected into the base region from the emitter region are collected to the collector region, so even if UCE continues to increase, the input characteristics of the transistor can hardly be changed.
Before introducing the output characteristics of transistor, it is necessary to introduce the concept of DC load line. The DC load line is the volt-ampere characteristic curve of collector load RC (output terminal resistance) when the transistor is working in static state, IC = (EC - UCE) / RC. When the transistor enters the off state, it is equivalent to the collector circuit entering the off state. At this time, UCE=0, the voltage on RC is equal to the power supply voltage EC. When the transistor enters the on state, ICM is the possible maximum value of the output current IC, that is, the maximum current flowing through RC. Mark these two points in a rectangular coordinate system with IC as the Y axis and UCE as the X axis, and draw a line segment, that is, the DC load line. The intersection of the DC load line with the Y-axis is called the saturation point, and the intersection with the X-axis is called the cut-off point. The slope of the DC load line is the resistance value of RC.
The intersection of the DC load line and the output characteristics curve of the transistor is called the quiescent operation point, or Q point. When the transistor works at the static operating point, no matter how the AC signal in the input signal changes, the transistor can be guaranteed to work in a stable amplification state (that is, the emitter junction is forward biased, and the collector junction is reverse biased), and no nonlinear distortion will occur. When selecting the Q point, try to stay away from the saturation region (to avoid saturation distortion) and cut-off region (to avoid cut-off distortion) of the transistor to obtain the best amplification effect.
Since the transistor mainly outputs through the reverse current of the collector junction J1, its output characteristic curve is very similar to the static characteristic curve under the reverse bias of the power diode, as shown in Figure 14. Compared with power diodes, transistors have three working states. In order to understand the relationship between the output current IC and the input current IB intuitively, the X axis can be extended to the left, and the left part of the X axis can be regarded as the positive X half axis of IB. The Q points are projected to the second quadrant, which divide the characteristic curve of IC and IB into four sections: 0, A, B and C. These sections correspond to working region of the transistor.
Cut-off Region (Section 0): When UBE≤UTO or IB=0, the emitter junction is in the off state. At this time, even if the collector junction is reverse biased (UBC<0), the transistor is still in the off state (in fact, there is a very small penetration current ICEO). Similarly, if the collector junction is in the off state (IC=0), even if the emitter junction is forward biased (UBE>0), the transistor will not be turned on. Therefore, the cut-off condition of the transistor is, IC * IB = 0.
Active Region (Section A): When the emitter junction is forward biased and greater than the threshold voltage (UBE>UTO>0), IB>0, if the collector junction is reverse biased (UBC≤0), the transistor works in the active region (amplification region). At this time, the value of IC has nothing to do with UCE, but is only affected by IB, and there is a linear relationship between IB and IC, IC = β * IB.
Saturation Region (Section B and Section C): As the base current increases, the number of holes in the base region decreases, and the carriers injected into the base region from the emitter region also decreases, and the depletion layer of the base region widens. When the saturation boundary is reached, the amplification capability of the transistor begins to weaken (β' = ΔIC / ΔIB < β), IB and IC no longer have a linear relationship, and the transistor starts to enter the quasi-saturation state (shallow saturation state, Section B). When the number of holes in the base region drops to a critical value, the potential of the base region is the same as the potential of the collector region, that is, the collector junction J1 is in zero bias (UBC=0), and the base current completely loses the amplification effect (β' = ΔIC / ΔIB =0), and the transistor enters the fully saturated state (deep saturation state, Section C).
When the transistor is shallowly saturated, the base current IB is small and the conduction voltage drop is large, that is, the equivalent resistance of the transistor is large, so it is easy to exit the saturation state. When the transistor is deeply saturated, the base current IB is large and the conduction voltage drop is small, that is, the equivalent resistance of the transistor is small, and as IB increases, the saturation of the transistor will continue to deepen, so it is difficult to exit the saturation state. In actual operation, when IB(sat) = EC / (β * ICM), it can be considered that the transistor has entered the deep saturation state, which is the saturation state in the usual meaning. Sometimes in order to accelerate the transistor into the deep saturation state, a base current that is several times IB(sat) is applied. It should be noted that the working status of the transistor is also affected by the output resistance RC -- the smaller the output resistance RC, the larger the saturation current IC, and the larger the saturation voltage drop UCE, and the larger the saturation trigger current IB. As the output resistance RC decreases, the saturation current IC will approach ICM, making the transistor easily burned. If the output resistance RC is close to 0, even if the transistor is burned out, it cannot enter the saturation state. Therefore, a larger output resistance can make the transistor more likely to enter the saturation state.
The increase in temperature will cause the intrinsic thermal excitation of the semiconductor, which will increase the carrier concentration inside the semiconductor and increase its conductivity. An increase in conductivity will cause an increase in leakage current, a decrease in threshold voltage, an increase in current gain and etc. Therefore, the input characteristic curve of the transistor will move to the right as the temperature rises, and the output characteristic curve of the transistor will move up as the temperature rises. The increase in temperature will also increase the possibility of the transistor thermal breakdown, so in actual use, sufficient heat dissipation conditions should be equipped to the transistor.
If the model of a transistor is known, its PCM parameters is also known. Through PCM=IC * UCE, PCM curve can be drawn. The ICM, UCEM, PCM curves can determine the safe operating area (SOA) of the transistor, in this area, the transistor can work stably without damage. The area outside the safe operating area is a hazardous area. In the hazardous area, the temperature of the transistor will increase significantly, making it more susceptible to thermal breakdown. Therefore, the transistor should be avoided to work in hazardous areas.
When the turn-on condition (UBE> UTO) is met, the transistor will be turned on. The turn-on process of the transistor is divided into the delay time td, the rise time tr, and the diffusion time ts.
The delay time td is the time taken from 10% IB1 to 10% IC1. This time period is the time required to charge the barrier capacitor.
The rise time tr is the time taken for IC to go from 10% IC1 to 90% IC1. During this time period, IC rose sharply.
The diffusion time ts is the time taken for IC to go from 90% IC1 to 100% IC1. This time period is the time required to charge the diffusion capacitor.
The calculation formula of the turn-on time: ton=td + tr + ts
When the cut-off condition (IB=0) is met, the transistor will be turned off. The turn-off process of the transistor is divided into the storage time ts, the fall time tf, and the tail time tt.
The storage time ts is the time taken from 90% IB1 to 90% IC1. This time period is the time required to remove the carriers stored in the base region during saturated conduction.
The fall time tf is the time taken for IC to fall from 90% IC1 to 10% IC1. During this time period, IC dropped sharply.
The tail time tt is the time taken for IC to fall from 10% IC1 to ICEO. This time period is the time required for the recombination of the remaining carriers.
The calculation formula of the turn-off time: toff = ts + tf + tt
Half-controlled device (also known as thyristor, or Silicon Controlled Rectifier SCR) is a bipolar device that can be turned on but not turned off through a control signal (gate trigger). The thyristor was born in 1956 and has a very wide range of applications in the 1960s and 1970s. However, with the birth of fully-controlled devices in the 1980s, the status of thyristors was gradually replaced. However, because the thyristor can withstand very large voltages and currents, and has a simple structure and reliable operation, it still retains an important position in large-capacity applications. The thyristor has three terminals. According to its shape, the thyristor can be divided into bolt type (usually the bolt is an anode, which can be tightly connected with the radiator and easy to install) and the flat type (the flat thyristor can be clamped by two radiators). In addition to gate triggering, the thyristor will also be turned on due to the following reasons: The anode voltage rises to a very high value and causes the avalanche effect, that is, the reverse biased PN junction in the middle is broken down; the anode voltage rise rate dv/dt is too high, that is, the junction capacitance effect of the PN junction; the junction temperature is high; light triggering. On the whole, only gate triggering is the most accurate, rapid and reliable control method. However, due to the development of semiconductor technology, modular thyristors are now common (click to view more thyristor modules).
The thyristor has a P-N-P-N four-layer structure, which has one more PN junction than transistor. The thyristor has three terminals -- anode A, cathode K, and gate G. The doping degree of the P-type semiconductor and the N-type semiconductor of the thyristor are different. The internal structure of the thyristor can be equivalent to two transistors V1 and V2, as shown in Figure 19, a. V1 is a PNP transistor (P+|N-|P). P+ region is the emitter region, N- region is the base region, and P region is the collector region. V2 is an NPN transistor (N+|P|N-). N+ region is the emitter region, P region is the base region, and N- region is collector region. Similar to the transistor, when using a thyristor, be careful not to connect the cathode and anode reversely to prevent the thyristor from being burned.
The equivalent working circuit of the thyristor is shown in Figure 19, b. V1 and V2 are equivalent transistors. The anode and cathode of the thyristor are connected to the output circuit, and the gate of the thyristor is connected to the input circuit. EA is the power supply in the output circuit, and EG is the power supply in the input circuit. R is the output resistance. IC1 is the collector current of V1, and IC2 is the collector current of V2. The current flowing through the anode is anode current IA, the current flowing through the cathode is cathode current IK, and the current flowing through the gate is gate current IG. α1 is the common base current gain of V1, and α2 is the common base current gain of V2. The idea of turning on the thyristor is similar to that of the transistor, that is, how to make the PN junction J2 generate a larger reverse current.
Cut-off State: When a forward bias voltage UAK is applied to the cathode and anode of the thyristor, and no voltage is applied to the gate, it is equivalent to that the collector and the base of V1 and V2 are open, and the thyristor is in the off state. Due to the effect of the forward bias voltage, the depletion layer of J1 and J3 becomes narrower, and the depletion layer of J2 becomes wider, so there is a reverse saturation current ICBO in J2 -- This current consists of two parts, one is the hole current ICBO1 (the common base current of V1), and the other is the free electron current ICBO2 (the common base current of V2). These two currents will flow through J1 and J3, forming the leakage current of the thyristor, which is slightly larger than the sum of the leakage currents of the two equivalent transistors. It should be noted that in the positive blocking state, α1 + α2 is very small.
Conduction State: When a forward bias is applied to the gate of the thyristor, the P+ region injects a large number of holes into the P base region -- one part of it enters the N+ region, making J3 forward conduction, and a large number of free electrons are injected into the P base region from the N + region, and the minority carrier concentration in the P base region increases, making ICBO2 increase; the other part of it enters the N- region, making the minority carrier concentration in the N- region increase, so ICBO1 increases. Both of these leakage currents will reduce the minority carriers in the N- region and narrow the depletion layer of J1. When a sufficiently large gate forward bias is given to make the depletion layer of J1 narrow to a certain extent, the dynamic balance of J1 is broken, and a large number of holes are injected into the N- region from the P+ region, which flows into the P region in the form of a reverse current and forms the current IC1, and then flows out of the thyristor from the N+ region in the form of a forward current; similarly, the free electrons in the N+ region flow from the P region to the P+ region, and form the current IC2. It should be noted that when IC1 and IC2 are not established, the base conductivity is very small, so α1 + α2 is very small. However, when IC1 and IC2 are established, due to the conductance modulation effect, the conductivity of the base region of V1 and V2 increases, which leads to the increase of IC1 and IC2. The positive feedback between these two currents makes α1 + α2 increase rapidly and approach 1, making the on-state voltage drop sharply drops, and the anode current IA sharply rises, and finally turning on the thyristor.
IC1 = α1 * IA + ICBO1, (10)
IC2 = α2 * IK + ICBO2, (11)
IK = IA + IG, (12)
IA = IC1 + IC2, (13)
IA = (α2 * IG + ICBO1 + ICBO2) /[1 - (α1 + α2) ]. (14)
It can be seen from Formula 14:
When α1+α2 approaches 0, IA will tend to leak current;
When α1+α2 approaches 1, IA will tend to infinity.
When the gate is open, the forward non-repetitive peak voltage UDSM (also known as the maximum off-state transient voltage) is the off-state peak voltage determined by the sharp bending point of the forward volt-ampere characteristic curve; the reverse non-repetitive peak voltage URSM (also known as the maximum reverse transient voltage) is the off-state peak voltage determined by the sharp bending point of the reverse volt-ampere characteristic curve.
The forward turning voltage UBO refers to the peak voltage that causes the thyristor to transit from the off state to the on state when a forward sin half-wave voltage is applied between the anode and the cathode of the thyristor and the gate is open at the rated junction temperature (100℃).
The reverse breakdown voltage UBR refers to the peak voltage that causes the reverse leakage current of the thyristor to increase sharply when a reverse sine half-wave voltage is applied between the anode and the cathode of the thyristor at the rated junction temperature (100℃).
The forward off-state repetitive peak voltage UDRM (also known as the off-state repetitive peak voltage) is the forward peak voltage allowed to be repeatedly applied to the device when the gate is open and the junction temperature is rated. The repetition rate is 50 times per second, and the duration of each time is not more than 10ms. Generally, UDRM is specified as 90% of UDSM. And UDRM should be 100V smaller than UBO.
The reverse off-state repetitive peak voltage URRM (also known as the reverse repetitive peak voltage) is the reverse peak voltage allowed to be repeatedly applied to the device when the gate is open and the junction temperature is rated. The repetition rate is 50 times per second, and the duration of each time is not more than 10ms. Generally, URRM is specified as 90% of URSM. URRM voltage should be lower than UBR.
UDRM and URRM will decrease with the increase of temperature. During testing and use, the temperature should be strictly regulated. Usually, the smaller one of UDRM and URRM is taken as the rated voltage of the thyristor.
The gate trigger voltage UGT refers to the minimum gate DC voltage required to transit the thyristor from the off state to the on state when a certain value of forward voltage is applied between the anode and the cathode of the thyristor under the specified ambient temperature. UGT is generally about 1.5V.
The forward average voltage drop UF (also known as on-state average voltage or on-state voltage drop) refers to the average value of the voltage drop between the anode and the cathode of the thyristor when the on-state current of the thyristor is the rated current under the specified ambient temperature and standard heat dissipation conditions. UF is usually 0.4-1.2V.
The on-state peak voltage UT refers to the transient peak voltage when the on-state current of the thyristor is a specified multiple of the rated current.
The rated on-state current IT refers to the maximum power frequency sine half-wave current value allowed to flow through the thyristor under the condition of the specified ambient temperature (40°C) and the specified cooling conditions, when the conduction angle is not less than 170°, the load is resistive, and the stable junction temperature does not exceed the rated junction temperature. The unidirectional thyristor uses the rated on-state average current IT(AV) as the rated current; the bidirectional thyristor uses the rated on-state effective current IT(RMS) as the rated current. If the current waveform is not a power frequency sine half-wave, although the thyristor is a semiconductor without the same volt-ampere characteristic curve as the resistance, a resistor with the same effective value (same heating effect) can be used as the reference to determine the rated current of the thyristor (1.5-2 times the equivalent calculation result of the resistance).
IDRM and IRRM are the corresponding leakage currents of UDRM and URRM respectively, generally less than 100μA.
The gate trigger current IGT refers to the minimum gate DC current required to transit the thyristor from the off state to the on state when a certain value of forward voltage is applied between the anode and the cathode of the thyristor under the specified ambient temperature. The IGT of ordinary thyristor is generally several milliamperes; the IGT of high-sensitivity thyristors is generally several micro-amperes.
The holding current IH is the minimum current required to keep the thyristor conducting. The IH is generally tens to hundreds of milliamperes. When the gate is triggered, even if the gate signal is removed, the thyristor is still on, and the thyristor can only be turned off by reducing the anode current. When the anode current is less than IH, the thyristor will be turned off. The higher the junction temperature, the smaller the IH, the less likely the thyristor is to be turned off.
The latching current IL refers to the minimum current required to keep the thyristor turned on after the thyristor has just turned from the off state to the on state and the gate signal is removed. For the same thyristor, IL is generally about 2-4 times of IH.
The inrush current ITSM refers to the non-repetitive maximum forward overload current when the junction temperature of the thyristor caused by the abnormal circuit exceeds the rated junction temperature during the half cycle of the power frequency sine wave. Normally, in a forward wave, the thyristor can withstand an overload current that is 6 times the rated current. During the life of the thyristor, if the limit of the number of surges is exceeded, the thyristor may be permanently damaged.
The forward turning current IBO refers to the peak current that can change the thyristor from the off state to the on state when the gate is open at the rated junction temperature (100℃).
The turn-on time tgt refers to the delay time for the thyristor to switch from the off state to the on state when enough trigger signals are applied. During the turn-on process, the output voltage UAK of the thyristor will gradually decrease to the on-state voltage drop UF, and the anode current IA will gradually increase to the rated on-state current IT.
The turn-off time tq refers to the time interval from when the on-state current of the thyristor drops to zero until the thyristor begins to withstand the specified off-state voltage. The tq of an ordinary thyristor is about several hundred milliseconds. The tq is not only related to the internal structure of the tube, but also related to the temperature, dv/dt, di/dt and etc. The tq can usually be reduced by increasing the reverse voltage. If the anode voltage is reapplied during the tq, the thyristor can be turned on again, but once the tq has passed, the thyristor will be turned off no matter how to increase the anode voltage (provided that the thyristor is not broken down).
See also:The tgt and the tq determine the operating frequency of the thyristor. For high operating frequency circuits, a thyristor with a small tq should be selected (if tq is small, tgt will be smaller). This parameter is the main difference between ordinary thyristors and fast thyristors.
The critical off-state voltage rise rate dv/dt refers to the maximum rise rate of the applied voltage of the thyristor from off-state to on-state conversion when the gate is open at the rated junction temperature. When the junction capacitor charging current is large, if the voltage rise rate is too large, the charging current will become large enough to cause the thyristor to turn on by mistake. The dv/dt of a small current thyristor (50-100A) is generally 225V/μs, and the dv/dt of a large current thyristor (above 200A) is generally greater than 50V/μs.
The critical on-state current rise rate di/dt refers to the maximum rise rate of the on-state current that the thyristor can withstand without damage when the gate is close at the rated junction temperature. The thyristor will produce a large power loss at the moment of turning on, and due to the limited conduction expansion speed, this loss is always concentrated in the cathode region near the gate. If the current rises too fast, even if the conducting current is not large, it will easily cause the thyristor to overheat locally, causing permanent damage to the gate, and causing the thyristor to be burned out. The larger the rated current of the thyristor, the more prominent this problem.
The static characteristics of the thyristor are the volt-ampere characteristics of the output current and output voltage, as shown in Figure 20. UAK is the voltage applied to the anode and the cathode of the thyristor, IA is the anode current, and IG is the trigger current.
Forward Blocking State: When IG = 0, even if UAK> 0, there is only a small forward leakage current. α1 + α2 is also very small. At this time, the thyristor is in the forward blocking state. But when UAK ≥ UBO or IA ≥ IBO, α1 + α2 approaches 1, and the thyristor will enter the forward conduction state.
Forward Conduction State: The common forward conduction situation is that under the condition of IG> 0, when UAK ≥ UGT, the conductivity of the thyristor base area increases significantly, α1 + α2 approaches 1, the current IA flowing through the thyristor will approach infinity (the actual value of IA is determined by the external circuit), and finally saturated conduction is achieved. Under the same external conditions, the larger IG, the smaller UGT. It should be noted that once the thyristor is turned on, the gate loses its control function. Only when the output current IA of the thyristor is reduced to a certain value close to 0 can the thyristor be turned off.
The reverse static characteristics of thyristors are similar to the reverse static characteristics of power diodes. When the thyristor is subjected to a reverse voltage, no matter whether the gate has a trigger current or not, the thyristor will not be turned on, and there is only a very small reverse leakage current. At this time, the thyristor is in a reverse blocking state. However, when the reverse voltage reaches the reverse breakdown voltage UBR, it will cause the thyristor avalanche breakdown.
When UAK1 is applied to the output terminal of the thyristor, the thyristor is in the off state at this time, and UAK is 100% UAK1. When UG ≥ UGT, it will take a period of time before the thyristor enters the conducting state. When the thyristor is turned on, its output terminal voltage UAK will maintain a very small value, that is, the on-state voltage drop.
The turn-on process of the thyristor is divided into the delay time td, the rise time tr, and the diffusion time ts.
The delay time td is the time required for IA to rise from the forward leakage current to 10% IA1, and UAK to decrease from 100% UAK1 to 90% UAK1. The delay time is generally 0.5-1.5μs. The delay time decreases as the gate current increases.
The rise time tr is the time required for IA to rise from 10% IA1 to 90% IA1, and UAK to fall from 90% UAK1 to 10% UAK1. The rise time is affected by the characteristics of the thyristor itself, the impedance of the external circuit, the temperature, the anode voltage and etc. The rise time is generally 0.5-3μs. By increasing IA, the delay time td and the rise time tr can be significantly shortened.
The diffusion time ts is the time required for IA to rise from 90% IA1 to 100% IA1, and UAK to drop from 10% UAK1 to the on-state voltage drop. The diffusion time depends on the cross-sectional area of the cathode.
Normally, when IA reaches 90% IA1, it can be considered that the thyristor has been turned on. Therefore, the formula for calculating the turn-on time is: tgt = td + tr.
By reducing UAK to 0 or applying a sufficiently large reverse voltage UAK2, IA is gradually reduced, and the thyristor is converted from the on state to the off state. The turn-off process of the thyristor is divided into the reverse blocking recovery time trr, and the forward blocking recovery time tgr.
During the turn-off process, due to the inductance of the external circuit, a reverse recovery current IR appears in the thyristor. When the IR gradually reaches the peak value IRP, a corresponding URP will be generated, and then the IR will rapidly decay. The time from IA falling to zero to IR falling to the reverse leakage current of the thyristor is called the reverse blocking recovery time trr.
The time from the end of the reverse recovery process to the complete recovery of the forward blocking capability of the thyristor is called the forward blocking recovery time tgr (or gate recovery time). During the tgr, because a small number of carriers are left on the PN junction near the gate, they can still trigger the positive feedback mechanism inside the thyristor. If a forward voltage is applied to the thyristor, the thyristor will conduct forward again. No gate trigger signal is required during this turn-on process.
The calculation formula of the turn-off time tq is: tq = trr + tgr.
By connecting multiple thyristors in series, the overall voltage capacity can be increased. In fact, it is not possible to multiply the withstand voltage value of a thyristor by the number to obtain the overall withstand voltage value. Instead, it should be added by the voltage actually borne by each thyristor. This is mainly because the voltage distributed on each thyristor is not uniform (divided into static uneven voltage and dynamic uneven voltage).
Static Uneven Voltage: Although the leakage current flowing through the series-connected thyristors is the same, because of their dispersion of static volt-ampere characteristics, the voltages assigned to each thyristor are not the same. In extreme cases, a certain thyristor may withstand all voltages, while other thyristors may only withstand very small voltages. The static uneven voltage can be reduced by selecting thyristors with very similar parameters and characteristics. The resistance equalization method can also be used to reduce static unevenness, that is, the thyristor is regarded as a high-resistance resistor (about 1 megohm), and each thyristor is connected in parallel with a low-resistance resistor to adjust the equivalence resistor of each parallel circuit. when their equivalent resistance values are very close, then the voltage distributed on each thyristor will also be very close.
Dynamic Uneven Voltage: The dynamic uneven voltage is caused by the difference in dynamic parameters and dynamic characteristics of the thyristor. By selecting thyristors whose dynamic parameters and characteristics are as consistent as possible, the dynamic uneven voltage can be reduced. It can also be triggered by a strong gate pulse to significantly reduce the difference in the turn-on time of the thyristor. It is also possible to use RC parallel branches for dynamic voltage equalization, that is, to absorb the over-voltage through the RC circuit, so that the voltage that each thyristor bears under dynamic conditions is very close.
By connecting multiple thyristors in parallel, the overall current capacity can be increased. Because of the different parameters and characteristics of each thyristor, it is also necessary to consider their uneven current distribution. By selecting thyristors with consistent characteristics and parameters as much as possible to reduce the dynamic uneven current and the static uneven current. It is also possible to reduce the dynamic uneven current by the current-sharing reactor (its loss is less than resistance). It is also possible to use strong gate pulse triggering to significantly reduce the difference in the turn-on time of the thyristors, so that each thyristor can be effectively triggered in a short time to achieve the purpose of dynamic current sharing. However, because the current capacity of the thyristor is getting larger and larger, it is usually unnecessary to operate the thyristor in parallel.
When thyristors need to be connected in series and in parallel at the same time, it is usually recommended to connect in series firstly and in parallel secondly to ensure that the parameters and characteristics of each thyristor are as consistent as possible.
The fast switching thyristor (FST) has excellent dynamic characteristics. Compared with ordinary thyristors, the FST has the advantages of short the turn-on time (generally 4-8μs), short turn-off time (generally 10-60μs), and large tolerance of dv/dt and di/dt. Ordinary thyristors can only work at a voltage of 50 Hz, while FST can work in circuits with higher frequencies (above 400 Hz). High-frequency thyristors (HFT) have shorter switching times and faster switching speeds than FST, and are suitable for working in high-frequency circuits (above 10kHz). Due to the high operating frequency, the heating effect of the switching loss of FST and HFT cannot be ignored, so their rated voltage and rated current are usually not high,
The bidirectional thyristor (also known as bidirectional triode thyristor, triode AC switch, TRIAC) can be considered as a pair of anti-parallel connected unidirectional thyristors (SCR). The bidirectional thyristor is a common core device in AC solid-state relays and modules (click to view more AC solid-state relays). The forward characteristic of the bidirectional thyristor is the same as that of the unidirectional thyristor, but its reverse characteristic is different from that of the unidirectional thyristor. The bidirectional thyristor does not have the reverse blocking ability. It can be clearly seen on the coordinate axis that the characteristic curve of the bidirectional thyristor is centrally symmetric. The bidirectional thyristor has a T1 pole (the main electrode connected to the P-type semiconductor material), a T2 pole (the main electrode connected to the N-type semiconductor material), and a gate G pole. The rated current of the bidirectional thyristor is the rated on-state effective current IT(RMS). There is no forward peak voltage and reverse peak voltage in the parameters of the bidirectional thyristor, but only the maximum peak voltage. The other parameters of the bidirectional thyristor are the same as the unidirectional thyristor.
The design idea of the reverse conducting thyristor (RCT) is similar to that of the bidirectional thyristor, but the reverse conducting thyristor uses a power diode for anti-parallel connection, so that the emitter junction of the anode and the cathode are both in the short-circuit state. Due to this special structure, the reverse conducting thyristor has the advantages of low on-state voltage, short turn-off time, high rated junction temperature, high voltage resistance, high temperature resistance, and etc. For example, the turn-off time (several microseconds) and power frequency (tens of kHz) of the reverse conducting thyristor are obviously better than that of the fast switch thyristor. Reverse conducting thyristor can be regarded as an organic combination of thyristor and freewheeling power diode, which can simplify circuit design, and is widely used in applications such as switching power supplies and UPS.
The light-triggered thyristor (also known as LTT, or light-controlled thyristor) is a thyristor that its gate region integrates a photoelectric power diode, and uses the strength of the light signal to replace the gate trigger current. Therefore, the light trigger is a kind of gate trigger. In order to improve the trigger sensitivity of the light-controlled thyristor, the gate region often adopts an amplified gate structure or a double-amplified gate structure. The light trigger ensures the electrical insulation between the main circuit and the control circuit, and can avoid the influence of electromagnetic interference. Low-power light-controlled thyristors are often used in electrical isolation to provide trigger signals for high-power thyristors. High-power light-controlled thyristors can ensure good insulation between the control circuit and the main circuit and are used in high-voltage power equipment (such as high-voltage direct current transmission).
The fully-controlled device (also called the self-shut-off device) is a device that can be controlled by a control signal to be turned on and to be turned off. Since the birth of fully-control devices in the 1980s, power electronics technology has entered a new era. There are many types of fully-controlled devices, most of which use composite structures, each with its own characteristics, which can meet the needs of various fields. Fully-controlled devices mainly include gate-off thyristors (GTO), bipolar junction transistors (including GTR, etc.), field effect transistors EFT (including MOSFET, JFET, etc.), and composite devices (including IGBT, MCT, SIT, SITH, IGCT, etc.). The following content will give a brief introduction to them.
The gate turn-off thyristor (GTO) is a derived device of the thyristor, which appeared soon after the advent of the thyristor. The ordinary thyristor (SCR) is the half-controlled device, and the gate signal no longer has any effect after the SCR is turned on. The GTO is the fully-controlled device that can be turned off by applying a negative pulse signal to the gate.
Although the structure of GTO is similar as the SCR, GTO has one more N+ buffer region, so the turn-on time of GTO is shorter than SCR, but at the same time its reverse blocking ability is weaker than SCR. GTO is a multi-unit power integrated device, which is composed of dozens or even hundreds of GTO units with a common anode. The cathodes and gates of these GTO units are connected in parallel inside the device. The cathode region of each GTO unit in this integrated structure is small, and the distance between the gate and the cathode is greatly shortened, which makes the P2 base region small in lateral resistance and can draw a larger current from the gate, which makes the collector of V1 is easy to be cut off, and finally makes the thyristor easy to be turned off. This integrated structure also makes the turn-on speed of GTO faster than that of SCR, and has stronger di/dt bearing capacity and overload bearing capacity -- SCR has only one unit, once the di/dt is too large, it is easy to be damaged by local overload and overheating; any local area of GTO is composed of multiple small GTO units, and the overload and overheating caused by the di/dt will be dispersed to these small GTO units. The volume of GTO is much smaller than SCR, so its capacity density is higher than SCR. At the same time, GTO does not require a commutation circuit, so it can be used in applications above 1kHz, while SCR can only be used in applications within 1kHz. But the reverse blocking ability of GTO is weaker than SCR, usually 20-30V. Because of these advantages, GTO is widely used in high-power applications above the megawatt level.
Generally, a power diode will be connected in anti-parallel to GTO to optimize its switching characteristics. In order to facilitate design and use, the power diode is normally integrated in GTO to form a reverse-conducting GTO, which is a bit similar to a reverse-conducting thyristor. The reverse-conducting GTO no longer has the ability to withstand reverse voltage. If it needs to withstand the reverse voltage, a power diode is required in series.
The turn-on principle of GTO is similar to the turn-on principle of thyristor. The N+ region contributes to the reduction of conductivity to speed up the positive feedback process. The α1 of GTO is designed to be small, which makes the saturation depth of GTO shallower than SCR, so it is easier to be cut off -- when SCR is on, α1 + α2 ≥ 1.15; when GTO is on, α1 + α2 ≈ 1.05, which is close to critical saturation. This design makes the equivalent transistor V2 more sensitive to the gate control signal, which makes V2 easy to be turned on and turned off, but at the same time, it also makes the on-state voltage drop of GTO increases. During the turn-off process, a sufficiently large negative current is passed into the gate to consume holes in the P base region and inject a large number of free electrons into the N- base region, with the holes of the P+ region injected into the base region decrease, IC1 gradually decreases, and IC2 also decreases. After a series of positive feedback process, when the anode current IA is less than the holding current IH, the GTO is cut off due to exiting saturation.
The calculation formula of GTO is the same as the calculation formula of thyristor.
IC1 = α1 * IA + ICBO1, (10)
IC2 = α2 * IK + ICBO2, (11)
IK = IA + IG, (12)
IA = IC1 + IC2, (13)
IA = (α2 * IG + ICBO1 + ICBO2) /[1 - (α1 + α2) ]. (14)
|IGRP|>(α1 + α2-1) * IATO/α2, (15)
βoff=IATO/|IGRP|. (16)
It can be seen from Formula 14:
When α1+α2 approaches 0, IA will tend to leak current;
When α1+α2 approaches 1, IA will tend to infinity.
Most of the parameters of GTO are the same as the main parameters of SCR.
The turn-on time ton is the sum of the delay time td and the rise time tr.
The turn-off time toff is the sum of the storage time ts and the fall time tf.
The maximum controllable anode current IATO is the rated current of GTO. If the anode current IA is greater than IATO, α1 + α2 cannot meet the condition slightly greater than 1, which will deepen the saturation so that GTO cannot be turned off normally.
The current turn-off gain βoff is the ratio of the maximum controllable anode current IATO to the peak gate negative pulse current IGRP. βoff=IATO/IGRP. Because the βoff of GTO is too small (usually 3-8), GTO can only be cut off by applying a very large negative current to the gate, which is equivalent to using a strong current to control a strong current. For example, a GTO with a rated current of 1000A requires a gate turn-off current of 300A. This main disadvantage limits the application of GTO.
The static characteristics of GTO are the same as the static characteristics of SCR, but the latching current IL (2A) of GTO is larger than that of SCR (100-500mA).
Similar to the turn-on process of SCR, when UAK=100% UAK1, and UG ≥ UGT, GTO will enter the conducting state, and its output terminal will generate a small on-state voltage drop. However, due to the multi-unit structure of GTO, its gate trigger current IGT is higher than that of SCR. The delay time td of GTO is about 1-2μs. The rise time tr of GTO increases with the increase of the on-state anode current.
Different with SCR, when a negative pulse voltage is applied to the gate to provide a large enough negative pulse current, GTO enters the turn-off process. The turn-off process of GTO is divided into the storage time ts, the fall time tf, and the tail time tt.
The storage time ts is the time required for IA to decrease from 100% IA1 to 90% IA1. When the turn-off signal UG2 is applied, the negative gate current rises rapidly from 0 to IGRP. The di/dt of the negative gate current depends on the circuit inductance and anode voltage. The negative gate current extracts the carriers stored during saturated conduction from the P base region of the GTO, so that the equivalent transistor V2 exits saturation. Within this period, GTO has not completely exited saturation, so UAK and IA remain unchanged.
The fall time tf is the time required for IA to decrease from 90% IA1 to 10% IA1. When the negative pulse current reaches IGRP, the anode current IA starts to drop rapidly, and the anode voltage UAK starts to rise. As the conductivity of the base region decreases, α1 + α2≤1, GTO begins to enter the turn-off state. Since the fall time tf is very short (about 2μs), and the fall rate of IA is very large, there will be a spike voltage at the output terminal of GTO.
The tail time tt is the time required for IA to decrease from 10% IA1 to 0. During this period, the remaining carriers will recombine, UAK gradually rises to UAK1, and IA gradually decreases to 0. Due to the snubber circuit, there will be a transient overshoot at the output terminal of GTO. If the voltage rise rate of UAK is too large, GTO may turn on again. By maintaining a proper gate negative pulse, the tail time tt can be effectively shortened.
Normally, the fall time tf is less than the storage time ts, and the storage time ts is less than the tail time tt, that is, tf <ts <tt. When IA drops to 10% IA1, it can be considered that GTO has been cut off, so the calculation formula for the turn-off time is: toff = ts + tf.
The giant transistor (GTR) was born in the 1970s, which is a kind of bipolar junction transistor that can withstand high voltage and high current, so GTR is also known as the power BJT. The switching time of GTR is much shorter than that of thyristor and GTO (usually a few microseconds), so GTR has a higher operating frequency -- the operating frequency of thyristor is generally tens of Hz; the operating frequency of GTO is generally several hundred Hz; the working frequency of GTR is generally 1-20kHz. The power capacity of GTR is very large, such as 1800V/800A/2kHz, 1400v/600A/5kHz, and 600V/3A/100kHz. Since GTR has the advantages of low saturation voltage, good switching characteristics, wide safe operating area, large power capacity, and strong self-shut-off capability and etc., so GTR replaces the place of thyristors, and is widely used in medium-capacity and medium-frequency fields, such as power supplies, motor control, and general inverters. However, the drive power required by the GTR is large, and the design of its drive circuit is very complex. At the same time, the GTR has poor surge current resistance and is easily damaged by second breakdown. Therefore, the GTR is gradually replaced by Power MOSFETs and IGBTs.
According to different structures, GTR can be divided into NPN type and PNP type (the following content takes NPN type GTR as an example). According to different structural forms, GTR can be divided into single-tube type GTR, Darlington type GTR (composite-tube type GTR) and GTR module. Single-tube type GTR has a low saturation voltage drop and a slightly fast switching speed, but the current gain β is small (usually around 10), the current capacity is small, and the drive power is large. It is generally used in small capacity inverter circuits. The Darlington type GTR has the advantages of large current gain β (usually up to tens or hundreds of times), large current capacity, small driving power consumption, but its saturation voltage drop is high, and the turn-off speed is slow.
The Darlington type GTR contains multiple units (each unit consists of a Darlington tube), and these units are connected in parallel through integrated circuit technology. The GTR module encapsulates two or more (4, 6, or even 7) single-tube type GTR or Darlington type GTR dies in a single tube case. The GTR tubes in the GTR module can form single bridge arm, single-phase bridge, three-phase bridge, and three-phase bridge with bleeder pipe. GTR module has the advantages of housing insulation, easy to design and install. At present, single-tube type GTR and non-modular Darlington type GTR are rarely used in inverter circuits, while GTR modules are still widely used.
The Darlington tube is also called a composite tube. By connecting two transistors in series to form an equivalent transistor -- the output signal of the first transistor is used as the base signal of the second transistor. The type of equivalent transistor is the same as that of the first transistor. Both transistors are operated in the amplifying region by an appropriate external voltage, so the current gain of this equivalent transistor is the product of the current gains of the two transistors. Darlington tubes are usually used to amplify very tiny signals in highly sensitive amplifying circuits, such as high-power switching circuits, power amplifiers and regulated power supplies.
The structure of GTR is similar to the structure of BJT. NPN type GTR is divided into emitter region, base region and collector region by two PN junctions (collector junction J1 and emitter junction J2) -- the emitter region has small area and high doping concentration; the base region has thin thickness (5-20μm) and low doping concentration; the collector region is divided into two parts, the N- collector drift region has a large area and low doping concentration, and the N+ substrate region has small area and high doping concentration.
The working principle and calculation formula of GTR are the same as the working principle and calculation formula of BJT. However, it should be noted that during the turn-on process, the N+ substrate region will inject a large number of free electrons into the N- drift region to increase the reverse current of J1.
Most of the parameters of GTR are the same as the main parameters of BJT.
When the voltage UCE applied to the output of the GTR exceeds the specified value, the GTR will be broken down, and this voltage is called the breakdown voltage BV. The breakdown voltage is not only related to the characteristics of GTR, but also related to the connection method of the external circuit, as shown in Figure 33.
BVCBO is the reverse breakdown voltage between the collector and the base when the emitter is open;
BVCEO is the breakdown voltage between the collector and the emitter when the base is open;
BVCER is the breakdown voltage between the collector and the emitter when the emitter and the base are connected by a resistor;
BVCES is the breakdown voltage between the collector and the emitter when the emitter and the base are short-circuited;
BVCEX is the breakdown voltage between the collector and the emitter when the emitter junction is reverse biased.
These breakdown voltages have the following relationship:
BVCBO>BVCEX>BVCES>BVCER>BVCEO
To ensure safety, in the actual process, it is recommended that the maximum working voltage be much lower than BVCEO.
The maximum collector-emitter voltage UCEM is the rated voltage of GTR. In order to ensure safe use, the maximum operating voltage UCEM will be lower than the breakdown voltage BVCEO.
The second breakdown mainly occurs in the case of high voltage and low current, and the corresponding second breakdown withstand capacity is called the second breakdown power PSB.
The static characteristics of GTR are similar to the static characteristics of BJT. The difference is that the low-current BJT will work in the amplification region, while the GTR only works in the saturation region and the cut-off region (that is, GTR has only on state and off state, and dose not have amplification state). It is because that when the GTR is working in the amplification state, its current is very large, and the power consumption is also large, which will make the GTR easily burned out due to severe heat. However, in the switching process (the process of switching back and forth from on state to off state), the GTR must cross the amplification region. Usually, this process will be very fast to avoid damage to the GTR.
The dynamic characteristics of GTR are similar to the dynamic characteristics of BJT.
By applying a forward base current IB1, GTR will enter the conducting state. The turn-on process is divided into the delay time td and the rise time tr.
The calculation formula of turn-on time: ton = td + tr
By turning off the base current IB, GTR can be cut off. And a reverse current can speed up the turn-off process. The turn-off process is divided into the storage time ts and the fall time tf.
The calculation formula of the turn-off time: toff=ts + tf
● Add an acceleration capacitor. By connecting a capacitor in parallel at both ends of the base resistance of the GTR, and using the feature that the capacitor voltage cannot change suddenly at the moment of commutation, to improve the switching characteristics of the GTR. Of course, a fast switch GTR tube with a relatively small junction capacitance can also be selected.
● Increase the drive speed. When the GTR is turned on, by providing a forward drive current with a certain amplitude and a steep front edge, td and tr can be reduced to accelerate the turn-on process of the GTR and shorten the turn-on time ton. However, the drive current cannot be too large, otherwise the diffusion time ts of the GTR will be increased due to oversaturation.
● Reduce the saturation depth. By reducing the saturation depth during turn-on process, the carriers stored on the base can be reduced to shorten the storage time ts of the GTR.
● Apply negative drive current IB2. When the GTR is turned off, by applying a negative driving current with a certain amplitude and overshoot, the speed of extracting carriers from the base can be accelerated to shorten the storage time ts of the GTR.
● Apply reverse base voltage UB2. When the GTR is turned off, the dissipation of the stored charge can be accelerated by increasing the reverse base voltage to shorten the storage time ts of the GTR. But the reverse base voltage should not be too large to avoid breakdown of the emitter junction.
First Breakdown: When the collector voltage exceeds the breakdown voltage BVCEO, the GTR will be broken down -- IC increases rapidly, and the output voltage UCE of GTR will remain at a certain value (that is, the maintenance voltage BVsus). When the first breakdown occurs, as long as the external circuit can limit the current after the breakdown, the GTR will not be damaged. After the collector voltage is reduced to less than BVCEO, the GTR will return to normal, and its operating characteristics will not change. Therefore, the first breakdown is reversible and cannot cause damage to the GTR.
Second Breakdown: When the first breakdown occurs, if current limiting measures are not taken immediately, the collector current IC will reach the critical point of second breakdown ISB, which will cause the local current density in the GTR to increase. The local area will heat up and further increasing the local current density. After a series of positive feedback, although the surface temperature of the GTR is not high, the local area inside it is damaged due to the high temperature, and a low resistance channel is formed between the collector and the emitter, and the collector voltage UCE drops, and the collector current IC rises sharply. This phenomenon is called second breakdown. The second breakdown is irreversible, it will cause permanent damage to the GTR and significantly degrade its working characteristics.
The maximum voltage UCEM, the maximum collector current ICM, the maximum dissipation power PCM, and the second breakdown power PSB constitute the second breakdown critical line. The area marked by the shadow in the second breakdown critical line is the safe operating area.
The field effect transistor (FET) is a unipolar device controlled by voltage. The working principle of FET is mainly to use an electric field to form a conductive channel in a semiconductor to control the conductivity of the semiconductor. FET can be divided into junction FET (JFET, also known as static induction transistor, SIT) and insulated gate FET.
The metal-oxide-semiconductor field effect transistor (MOSFET) is an insulated gate field effect transistor made of metal, oxide and semiconductor. MOSFET is a very important power electronic device. MOSFET has high input impedance (about 107-1015Ω), low noise, low power consumption, large dynamic range, good temperature characteristics, easy to integrate, no second breakdown phenomenon, simple drive circuit, small drive power, fast switching speed, high operating frequency, good thermal stability (better than GTR), wide safe operating area, and the gate bias can be positive or negative or zero. Because of these advantages, MOSFETs are widely used in circuits, such as signal amplification, impedance conversion, variable resistors, constant current sources, electronic switches, and etc. Compared with other power electronic devices, due to the small current capacity and the low withstand voltage of MOSFET, a multi-unit integrated mechanism is usually used to form the Power MOSFET with high power capacity, which is mostly used in high-frequency power electronic equipment below 1kW. MOSFETs are mainly used in DC solid state relays (click to view more DC solid state relays).
MOSFET has four terminals -- gate (G), drain (D), source (S), and body (B). The main structure of the MOSFET is a substrate (also called body or base), whose main material is silicon. There is an oxide insulating layer (also called a gate oxide layer or a gate insulating layer) between the source and the drain, whose material is normally SiO2. The gate is on the gate insulating layer, whose material is used to be aluminum (aluminum gate), but now is polysilicon (poly-silicon gate). The internal resistance of the gate of the MOSFET is extremely high (up to several hundred megaohms), so there is no conduction between the gate, the source, the drain, and the substrate. In actual use, a power diode is usually connected in parallel between the drain and source of the MOSFET to avoid the instantaneous reverse current in the circuit from breaking down the MOSFET. Generally, the power diode is directly integrated into the MOSFET.
According to the material of the substrate, MOSFET can be divided into N-MOSFET (or N-MOS) and P-MOSFET (or P-MOS). N-MOS uses a P-type substrate, and its source must be connected to the lowest potential of the circuit; P-MOS uses an N-type substrate, and its source must be connected to the highest potential of the circuit.
According to whether there is a conductive channel between the source and drain when the gate voltage is zero, MOSFETs can be divided into enhancement type MOSFET and depletion type MOSFET.
According to whether the source and drain are on the same plane, the structure of the MOSFET can be divided into a vertical structure and a horizontal structure.
There are many branch types and derivative devices of MOSFET, which cannot be listed here. The following is an introduction to LDMOSFET, CMOS and VDMOSFET.
The source (S) and drain (D) of the LDMOSFET are on the same plane, so a separate body (B) is required. Take N-channel LDMOSFET as an example (as shown in Figure 39). Its substrate is a P- substrate region, and the carriers involved in conduction are free electrons. There are two highly doped N+ regions in the P- substate region (P- Sub), one of which is called the source region and the other is called the drain region. These two regions are exactly the same, and theoretically it is no problem to swap them. Since the source (S) and drain (D) are on the same plane, the threshold voltage shift caused by the body effect is prone to occur. Therefore, by adding a high-doped P+ region to the P- substrate region, the body (B) is connected to the source (S) from the P+ region, so that the potential of the body is equal to the potential of the source, and the body effect can be avoided. The conductive channel in the LDMOSFET is formed in the substrate -- when the gate voltage of the enhanced N-channel LDMOSFET is 0, there is no channel between the source and the drain, as shown in Figure 39, a; when the gate voltage of the depletion N-channel LDMOSFET is 0, there is a channel between the source and the drain, as shown in Figure 39, b. By increasing the lateral dimension (horizontal area), the channel length of LDMOSFET increases. However, the increase of the horizontal area will lead to the increase of the on-resistance of LDMOSFET.
The distance between the source region and the drain region is called the conductive channel length. Ldrawn = Leff + 2 * LD. Ldrawn is the total channel length, Leff is the effective channel length, and LD is the lateral diffusion length. The distance from the gate to the bottom of the conductive channel is called the conductive channel width. The length (L) and width (W) of the conductive channel determine the channel resistance of the MOSFET.
Body effect refers to that the width of the depletion region at the conductive channel is wider after the substrate voltage is lower than the source voltage, which will result in a higher threshold voltage.
CMOS is one of the most commonly used processes in integrated circuits. It can integrate multiple N-MOS and P-MOS on a single silicon chip. P-MOS is directly embedded in N-MOS, so the substrate of P-MOS is also called N- Well. CMOS has the advantages of high efficiency, low power consumption, high impedance, and the ability to process complex logic.
The vertical double-diffused MOSFET (VDMOSFET) is the most common power MOSFET (Figure 41 shows the enhanced N-channel VDMOSFET). The source and drain of the VDMOSFET are not on the same plane, so a separate body (B) is not required. By increasing the vertical dimension (vertical area) without increasing the horizontal area, the channel length of VDMOSFET increases. Therefore, for MOSFETs with the same on-resistance, VDMOSFET has higher current withstand capabilities than LDMOSFET. Because there is an extra layer of N- epitaxial region with low doping concentration between the P region and the N region of VDMOSFET, which increase the equivalent resistance of the VDMOSFET, so VDMOSFET can withstand a higher voltage. Although due to the conductance modulation effect, when the current flowing through the PN junction gradually increases, the internal resistance of the N- epitaxial region will decrease, but the voltage withstand capability of the VDMOSFET is still greater than that of the LDMOSFET. In addition, the N region and the P region form a parasitic power diode, which can effectively protect the VDMOSFET from reverse voltage breakdown. Therefore, most power MOSFETs adopt a vertical structure to obtain better withstand voltage and current capability.
According to the shape of the groove gate, VDMOSFET can be divided into VVDMOS (also known as v-groove vertical double-diffused MOSFET, or v-groove vertical channel double diffused MOSFET, VMOS) and UVDMOS (also known as u-groove vertical double-diffused MOSFET, or u-groove vertical channel double diffused MOSFET, UMOS). UMOS has a better withstand voltage capability than VMOS. Since VMOS is very widely used in the field of power electronics, generally power MOSFET refers to VMOS.
The working principle of MOSFET is to change the conductivity inside the semiconductor by applying an external electric field. The working principle of P-MOSFET is exactly the same as that of N-MOSFET. In addition, the working principles of VDMOSFET and LDMOSFET are similar, but the lateral structure facilitates clear observation of current changes inside the MOSFET. So, the following takes the N-channel LDMOSFET as example to introduce the working principle of MOSFET.
When the gate voltage is not applied (UGS=0), there is no conductive channel between the source region and the drain region of the enhancement type MOSFET, and the conductive channel will be formed only after a certain gate voltage is applied. Therefore, the enhancement type MOSFET is a normally closed (NC) device.
Accumulation Layer Stage: When UGS<0, an electric field perpendicular to the semiconductor surface will be generated at the gate, which will attract holes in the P- region to the bottom of the insulating layer and repel the free electrons in the P- region, and then form an accumulation layer. Even if a positive voltage is applied between the source and the drain, current cannot flow directly from the drain to the source (the leakage current is not considered), so the MOSFET is in the off state.
Depletion Layer Stage: When 0<UGS<UT, an electric field perpendicular to the semiconductor surface will be generated at the gate, which will attract free electrons in the P- region to the bottom of the insulating layer and repel the holes in the P- region, and then form a depletion layer. However, because the free electrons accumulated at the bottom of the insulating layer are too few to form a conductive channel, the MOSFET is still in the off state.
Inversion Layer Stage: When UGS≥UT, the free electrons accumulated at the bottom of the insulating layer is enough to form an N type narrow layer (that is, N type conductive channel). Since the type of the N type conductive channel is opposite to that of the P- region, it is also called an inversion layer. The existence of the conductive channel allows current to flow from the drain region to the source region, so the MOSFET is in the on state. With the increase of UGS, the stronger the electric field formed by the gate, the more free electrons are attracted to the bottom of the insulating layer, the wider the conductive channel, the smaller the channel resistance, and the larger the drain current ID. It should be noted that the voltage drop generated by the drain current ID along the channel makes the voltage between each point in the channel and the gate no longer equal -- the closer to the source region, the greater the voltage drop and the wider the channel; the closer to the drain region, the smaller the voltage drop and the narrower the channel. The voltage drop at the end closest to the drain region is the smallest (its value is UGD= UGS - UDS), the channel is the narrowest. And as UDS increases, the channel near the drain region becomes narrower and narrower.
Pinch-off Region Stage: When the MOSFET is turned on, if UDS keeps increasing until UGD = UGS - UDS = UT, the channel width of the end closest to the drain region becomes 0, and a strong inversion channel cannot be formed, that is, the channel is pinched off. The point on the channel with a width of 0 is called the pinch-off point, and the depletion region between the pinch-off point and the drain region is called the pinch-off region. If UDS continue to increase, the pinch-off point will move to the source region. Since the increase of UDS is almost applied to the pinch-off region, the pinch-off region will also continue to expand. After the channel is pinched off, even if the drain voltage continues to increase, the drain current remains at a constant value (that is, the saturated drain current ID(sat)), and the MOSFET is in a saturation state. The saturated drain current ID(sat) is not affected by UDS, but is determined by UGS -- ID(sat) has a relationship with the square of UGS, which is the MOSFET square-law transfer characteristics.
When the gate voltage is not applied (UGS=0), there is a conductive channel between the source region and the drain region of the depletion MOSFET. When a forward gate voltage is applied, the width of the conductive channel will increase. When the reverse gate voltage is applied, the width of the conductive channel will decrease. When the reverse gate voltage reaches a certain value, the conductive channel disappears. Therefore, the depletion N-MOSFET is a normally opened (NO) device.
Conduction State: When UGS≥0, the electric field generated on the gate will attract more electrons to the N channel and repel the holes in the N channel. As UGS increases, the channel becomes wider, the channel resistance becomes smaller, and ID increases.
Cut-off State: When UGS<0, the electric field generated on the gate will attract more holes to the N channel and repel the electrons in the N channel, making the channel narrower and the channel resistance larger. When UGS reaches the pinch-off voltage UPO, the electrons in the N channel are depleted, the conductive channel disappears, and ID tends to zero.
Most of the parameters of MOSFET are the same as the main parameters of BJT.
Generally speaking, the ratio of the input current to the input voltage is called admittance. Due to the transfer characteristics of MOSFET, its gate-source input voltage UGS does not affect the change of gate-source input current IGS, but affects the change of drain current iD, so the ratio of the drain current ID to the gate-source voltage UGS is called transconductance.
There are three internal parasitic capacitances inside the MOSFET: The gate-source parasitic capacitance CGS, the gate-drain parasitic capacitance CGD (also known as Miller capacitance), and the drain-source parasitic capacitance CDS. These parasitic capacitances will affect the dynamic characteristics of the MOSFET -- if the parasitic capacitance is small, the switching current and the driving power are small, and the switching speed is fast; if the parasitic capacitance is large, the switching current and the driving power are large, and the switching speed is slow. MOSFET manufacturers usually give the capacitance parameters of the MOSFET, which meet the following calculation formulas:
Input capacitance of MOSFET: Ciss = CGS + CGD
Output capacitance of MOSFET: Coss = CDS + CGD
Reverse transfer capacitance of MOSFET: Crss = CGD
The drain-source on-resistance RDS(on) refers to the on-resistance between the drain and source of the MOSFET when it is turned on under certain conditions. RDS(on) is determined by the parasitic resistance of MOSFET, RDS(on) = RCS + RN+ + RCH + RA + RJFET + RD + RSUB + RCD. RCS is the contact resistance between the N+ source region and the source electrode; RN+ is the resistance between the N+ source region and the channel; RCH is the resistance of the channel; RA Is the resistance of the accumulation layer; RJFET is the resistance of the equivalent JFET; RD is the resistance of the drift region; RSUB is the resistance of the substrate region; RCD is the contact resistance between the substrate region and the drain electrode. The drain-source on-resistance RDS(on) is affected by the MOSFET junction temperature and the gate-source voltage (driving voltage) UGS. The higher the junction temperature, the larger RDS(on); on the contrary, the smaller RDS(on). The higher the gate-source voltage, the smaller RDS(on); on the contrary, the larger RDS(on).
The maximum drain current IDM is the rated current of the MOSFET. The maximum drain current refers to the drain current that enables the MOSFET to reach the highest junction temperature when the case temperature is at a certain value. The maximum drain current is not only related to the structure of the MOSFET, but also related to the packaging method of the MOSFET and the ambient temperature.
The maximum drain-source voltage BVDSS is the rated voltage of the MOSFET. The maximum drain-source voltage refers to the maximum voltage that can be applied when the MOSFET drain and source do not undergo avalanche breakdown when the ambient temperature is 25°C. In actual operation, the measured drain-source voltage value when the drain current is 250μA is usually taken as the maximum drain-source voltage BVDSS.
The maximum gate-source voltage (also known as the maximum driving voltage) refers to the maximum input voltage that can cause permanent damage to the gate insulating layer of the MOSFET in a very short time. It is generally recommended that the driving voltage should not exceed ±20V.
The switching time of MOSFET is between 10-100μs, and the operating frequency can reach above 100kHz (even up to several MHz), which is the highest among major power electronic devices. Although the MOSFET hardly needs input current when it is static, it will charge and discharge the parasitic capacitance when it is dynamic (switching process), so a certain drive power is still required. The higher the switching frequency of the MOSFET, the greater the drive power required.
Take the enhanced N-MOSFET as an example. VDD is the output power source of MOSFET; UP is the driving signal source of MOSFET; UGS is the voltage between the gate and source; UDS is the output voltage drop of the MOSFET; ID is the drain current; RS is the internal resistance of the drive circuit; RG is the gate internal resistance; RL is the drain load; RF is the detection resistance, used to detect the drain current.
Due to the transfer characteristics of the MOSFET, the voltage change between the gate and source of the MOSFET will not affect the current between the gate and source, but it will affect the drain current ID. When ID is large, the relationship between ID and UGS is approximately linear, and its slope is the transconductance Gfs of the MOSFET.
The static output characteristic curve of MOSFET is similar to the static output characteristic curve of GTR. The static output characteristic of MOSFET can be divided into cut-off region, saturation region and non-saturation region. MOSFET usually only works in the switching state (that is, fast switching back and forth between the cut-off region and the non-saturation region) to prevent the MOSFET from being burned out due to excessive power consumption when working in the saturation region.
Cut-off Region: Similar to the cut-off region of BJT. Although UDS is very high, the drain current ID=0 (the leakage current is not considered).
Saturation Region: Similar to the active region of BJT. ID is not affected by UDS, but increases with the increase of UGS. In the saturation region, the voltage and current that the MOSFET bears are large, so its power consumption is very large.
Non-saturation Region: Similar to the saturation region of BJT. ID increases with the increase of UDS. In the non-saturation, the MOSFET bears a large current and a very small voltage, so its power consumption is small. Because MOSFET behaves as a voltage-controlled resistor in the non-saturation region, so this region is also known as ohmic region or variable resistance region.
In the switching process of MOSFET, the influence of the internal parasitic capacitance on its switching time cannot be ignored.
In order to switch the MOSFET to the on-state, a steep input power source UP1 must be applied to its gate. Due to the existence of the internal resistance RS of the driving circuit and the gate-source parasitic capacitance CGS, the gate voltage UGS of the MOSFET cannot form a pulse waveform as steep as UP1, but rises with a certain slope. When the driving current starts to charge CGS so that UGS reaches UT, the MOSFET enters the on state, and the drain current ID starts to rise. When CGS is fully charged, UGS is maintained at UGS1, and ID is maintained at ID1. At this time, the drain-source parasitic capacitance CDS starts to discharge, and UDS starts decrease. When UDS reaches the minimum value, the driving current starts to charge the gate-drain parasitic capacitance CGD, and UGS rises again until it remains at UGS2. Generally, the time from UGS rising to 10% UGS2 to ID rising to 10% ID1 is called the turn-on delay time td(on). The time taken for ID to rise from 10% ID1 to 90% ID1 is called the rise time tr. The turn-on time ton of the MOSFET is the sum of the turn-on delay time td(on) and the rise time tr.
The calculation formula of the turn-on time: ton = td(on) + tr
When the driving pulse signal UP1 is removed, because of RS and CGD, UGS decreases with a certain slope. When CGD is discharged, UGS remains at UGS1, at this time, CDS starts to charge, and UDS starts to rise. When CDS is fully charged, UDS remains at 100% UDS1, at this time, CGS starts to discharge, and UGS drops again. When UGS drops below UT, the MOSFET enters the off state, and ID drops to 0. Since the MOSFET does not have a minority carrier storage effect, its turn-off process is very fast (about tens of nanoseconds). Generally, the time from 90% UGS2 to 90% ID1 is called the turn-off delay time td(off). The time it takes for ID to fall from 90% ID1 to 10% ID1 is called the fall time tf. The turn-off time ton of the MOSFET is the sum of the turn-off delay time td(off) and the fall time tf.
The calculation formula of the turn-off time: toff = td(off) + tf
● Using a drive circuit with low internal resistance and inductance can speed up the turn-on process of the MOSFET.
● Improve the MOSFET's ability to charge and discharge the parasitic capacitance during the turn-on and turn-off process, which can effectively reduce the delay time, so that no transient error occurs.
Because the switching speed of MOSFET is very fast, and its dynamic characteristics have a certain degree of dispersion, it is impossible to adopt common voltage equalization schemes. If MOSFETs connected in series, during the switching process, the working state of each MOSFET is different, and its withstand voltage capability cannot be kept the same, so it is easy to cause the MOSFET with low withstand voltage to burn out due to overvoltage. Therefore, MOSFETs are not suitable for series connection.
The on-resistance of the MOSFET has a positive temperature coefficient -- the higher the temperature, the greater the on-resistance. In practical applications, multiple MOSFETs can be used in parallel, for example, the output circuit of the inverter welding machine will connect dozens of MOSFETs in parallel to increase its current capacity. When one of the MOSFETs passes too much current, the greater the on-resistance, the more current will flow into other MOSFETs with smaller on-resistance, and finally the current flowing in each MOSFET tends to be balanced. The following methods can effectively reduce the phenomenon of uneven current: Select MOSFETs with small parameter error; the wiring and layout of the circuit should be symmetrical; the line loss (the resistance of the wire itself will consume current) and the inductance and capacitance of the wire under high frequency should be considered. It should be noted that connecting a small inductor in the source circuit can act as a current-sharing reactor, which can effectively reduce the dynamic uneven current, but it is invalid for the static uneven current.
The insulated-gate bipolar transistor (IGBT, or IGT) is a composite Bi-MOS device with the high input impedance of Power MOSFET and the high current capacity of BJT. IGBT is widely used in many fields, such as converters, inverters, pulse width modulation systems (PWM), uninterruptible power supplies (UPS), switching mode power supplies (SMPS), resonant converters, industrial motors, new energy vehicles, etc. IGBT has the advantages of high input impedance, low noise, fast switching speed, simple driving circuit, low driving power, low on-state voltage, low switching loss, wide safe operating area, small size, high current density, high current capacity, high endurance voltage, strong resistance to pulse current impact, and no second breakdown. Compared to Power MOSFET, IGBT has the disadvantages of slow switching speed and easy to latch-up. IGBT is mainly used in fields where the withstand voltage is above 600V, the current is above 10A, and the frequency is above 1kHz.
The structure of IGBT is very similar to that of Power MOSFET -- the gate (G) of the IGBT corresponds to the gate (G) of the MOSFET; the emitter (E) of the IGBT corresponds to the source (S) of the MOSFET; the collector (C) of the IGBT corresponds to the drain (D) of MOSFET. However, IGBT has a P+ injection layer in substrate region which makes the IGBT become a P-N-P-N structure like a thyristor. According to whether it contains N+ buffer layer, IGBT can be divided into Punch-Through type (PT) and Non-Punch-Through type (NPT). PT type IGBT has the advantages of low switching loss, low on-state loss, and large current capacity, but its temperature characteristic is not as good as NPT, and it is not suitable for parallel connection. The forward breakdown voltage of PT type IGBT is higher than the reverse breakdown voltage, so it is more suitable for DC circuits; the forward breakdown voltage of NPT type IGBT is the same as the reverse breakdown voltage, so it is more suitable for AC circuits. Since the switching speed of P-Channel IGBT is 2-3 times slower than that of N-Channel IGBT, the safe operating area (SOA) of P-Channel IGBT is smaller than that of N-Channel IGBT, and the cost of P-Channel IGBT is higher than that of N-Channel IGBT, so P-Channel IGBT is rare in actual use. The following mainly introduces PT type N-Channel IGBT.
The equivalent circuit diagram of the PT type N-Channel IGBT is a circuit that composed of a parasitic enhancement N-MOSFET, a parasitic JFET, a parasitic PNP transistor V1 (P+ N- P), and a parasitic NPN transistor V2 (N+ P N-), and an equivalent extended resistance R2, as shown in Figure 51, a. In this circuit, V1 is the main output channel; V2 is formed with the formation of MOSFT; JFET is mainly formed by the N- drift region; R2 is mainly formed by the equivalent resistance of the P base region of V2. The parasitic JFET can be further simplified as the equivalent modulation resistance R1 which is mainly formed by the equivalent resistance of the N- drift region. And then the equivalent circuit diagram of the IGBT can be simplified as shown in Figure 51, b. The four-layer semiconductor structure (P-N-P-N) of the IGBT can be regarded as a parasitic thyristor SCR, and the equivalent circuit diagram of the IGBT can be further simplified as shown in Figure 51, c. If the IGBT is regarded as a MOSFET with large current switching capability, the P+ and N+ regions can be regarded as a power diode VD1, and the equivalent circuit diagram of the IGBT can be further simplified as shown in Figure 51, d. The simplified equivalent circuit diagram helps to intuitively understand the working principle of the IGBT.
In simple terms, the working principle of IGBT is a voltage-drive thyristor.
Forward Blocking State: When a forward voltage is applied to the IGBT and the gate and emitter are short-circuited, the IGBT will enter a forward blocking state. At this time, the PN junctions J1 and J3 are forward biased, and the PN junction J2 is reverse bias. The reverse voltage makes the depletion layer on both sides of J2 extend to the P base region and the N-drift region.
Reverse Blocking State: When a reverse voltage is applied to the IGBT, the PN junction J1 is reverse biased, and the reverse voltage makes the depletion layer of J1 extend to the N- buffer region. By increasing the width of the N- buffer region, the reverse blocking capability of the IGBT can be improved, but it will also increase the forward voltage drop of the IGBT. The reverse withstand voltage of IGBT is usually only a few tens of volts, so in order to prevent the IGBT from working in the reverse blocking state, a FRED is connected in anti-parallel to the IGBT. Of course, for convenience, IGBT and FRED will be packaged together to form an reverse-conducting IGBT module.
Conduction State: When a forward voltage is applied to the IGBT and a certain voltage is applied to the gate, the P base region will form a N sub-channel region, allowing electrons to be transferred from the N+ emitter region to the N- drift region. This electrons flow will reduce the potential of the N base region and provide the base current IB1 for V1. If the voltage drop generated by this electrons flow is about 0.7V, then the PN junction J1 will be forward biased, and the IGBT start to be turned on. The N- drift region of IGBT is very wide and the doping concentration is low, so the N- drift region has a very low conductivity. When the IGBT is working at high current, due to the conductance modulation effect, the carrier concentration of the N- base region increases and its conductivity increases, which will reduce the saturation voltage between the collector and the emitter and the total on-state power consumption of the IGBT. When there are electron current IN and hole current IP, the IGBT is completely turned on. When the IGBT is in the on state, its collector current should be limited to avoid latch-up effect.
Cut-off State: When the gate voltage UGE is lower than the threshold voltage UT or a reverse bias voltage is applied to the gate, the N sub-channel disappears, the base current in the IGBT is cut off, and then IN and IP disappear, IGBT enters the cut-off state. However, due to the minority carrier effect, the IGBT output current will not be reduced to zero immediately, but a tail current will be generated like BJT, whose characteristics are related to UCE, IC and TC. The minority carrier effect will increase the switching time and switching loss of the IGBT.
Usually, R2 will short-circuit the base and emitter of V2 to prevent V2 from working. When the current flowing through R2 is too large, so that the forward voltage drop on R2 is sufficient to provide the trigger current IB2 for V2, then V1 and V2 will form a equivalent thyristor SCR (P+ N- P N+). Due to the positive feedback mechanism inside the equivalent thyristor, V1 and V2 will enter a deep saturation state, so the channel of IGBT is difficult to be shut off by the gate voltage, and IGBT can be shut off only if a very large reverse voltage is applied. This phenomenon is called the latch-up effect, which can be divided into static latch-up and dynamic latch-up. The static latch-up is caused by the excessive collector current IC when the equivalent thyristor is completely turned on. The dynamic latch-up is caused by the large displacement current caused by excessive di/dt and dv/dt during the switching process of IGBT. The collector current that causes the dynamic latch-up is smaller than that of the static latch-up. The latch-up effect makes V1 and V2 form a Darlington structure, so IC and power consumption of the IGBT will increase significantly, which will damage the IGBT. The following measures are usually taken to avoid the latch-up effect:
● Reduce R2 by changing the internal structure and doping of the IGBT to prevent V2 from turning on.
● By optimizing the width and doping of the N- buffer layer to reduce the current gain α of V1 (generally less than 0.5) to suppress the work of V2.
Most of the parameters of IGBT are the same as the main parameters of MOSFET.
The latching current IL refers to the value of the collector current that will cause the latch-up effect of the IGBT. The latching current IL is usually more than 5 times of the ICM (direct current). The latching current IL used to be one of the main reasons for limiting the current capacity of IGBT. However, with the development of technology, there is no need to consider the static latch-up when designing and using IGBT, but it is still necessary to the prevent dynamic latch-up.
The static input characteristic curve of IGBT is similar to the static input characteristic curve of MOSFET.
The static output characteristics of IGBT can be divided into forward blocking region, active region, saturation region, and reverse blocking region. IGBT usually only works in the switching state (that is, fast switching back and forth between the forward blocking region and the saturation region) to prevent the IGBT from being burned out due to excessive power consumption when working in the active region.
Forward Blocking Region: Similar to the cut-off region of BJT. When UGE < UT, the internal MOS channel of the IGBT is pinched off, and there is a leakage current ICEO between the collector and the emitter.
Active Region: Similar to the active region (amplification region) of BJT. When UGE ≥ UT and UCE > UGE - UT, IGBT works in the active region and will produce a 0.7V on-state voltage drop. In the active region, the electron current IN flowing into the N base region is controlled by the gate voltage UGE, which limits the base current IB1 of V1, and then the hole current IP is limited, so the collector current IC will enter a saturation state (similar to the saturation state of a MOSFET). In the active region, the voltage and current that the IGBT bears are very large, and the power consumption of the IGBT is also very large, so IGBT should cross this region as soon as possible.
Saturation Region: Similar to the saturation region of BJT. The saturation region of IGBT is also known as ohmic region or variable resistance region. When UGE ≥ UT, and UCE ≤ UGE - UT, the collector current IC is no longer controlled by the gate voltage UGE, but determined by the external circuit.
Reverse Blocking Region: Similar to the reverse blocking state of power diode.
The saturation voltage drop after the IGBT is completely turned on mainly depends on the conductance modulation, while the turn-on voltage drop of the MOSFET mainly depends on the drain current (resistance characteristic). Therefore, the saturation region of MOSFET refers to current saturation, and the saturation region of IGBT refers to voltage saturation.
The dynamic characteristics of IGBT are similar to the combination of MOSFET and BJT.
The turn-on process of IGBT is similar to that of MOSFET. Generally, the time taken from UGS rising to 10% UGS1 to IC rising to 10% IC1 is called the turn-on delay time td(on). The time taken for IC to rise from 10% IC1 to 90% IC1 is called the rise time tr. The turn-on time ton of the IGBT is the sum of td(on) and tr. However, it should be noted that the falling process of UCE is divided into two stages -- the tfv1 stage is the stage when the equivalent MOSFET works alone; the tfv2 stage is the stage when the equivalent MOSFET and the equivalent BJT work together.
The calculation formula of the turn-on time: ton = td(on) + tr
The turn-off process of IGBT is similar to that of MOSFET and BJT. Generally, the time taken from UGS falling to 90% UGS1 to IC falling to 90% IC1 is called the turn-off delay time td(off). The time taken for IC to fall from 90% IC1 to 10% IC1 is called the fall time tf. The turn-off time toff of the MOSFET is the sum of td(off) and tf. However, it should be noted that the falling process of IC is divided into two stages -- the tfi1 stage is the stage when the equivalent MOSFET works alone; the tfi2 stage is the stage when the equivalent MOSFET and the equivalent BJT work together. The tail time tt is the time required for the reverse recovery current to disappear. Due to the holes injected into the P+ collector region are recombined in P+ collector region, so the residual current is reduced and the tf is shortened.
The calculation formula of the turn-off time: toff = td(off) + tf = td(off) + tfi1 + tfi2
Positive-biased safe operating area (FBSOA): Determined by ICM, UCEM and PCM.
Reverse-biased safe operating area (RBSOA): Determined by ICM, UCEM and dUCE/dt.
Similar to MOSFET, IGBT is not suitable for series connection.
The temperature characteristic of the on-resistance RON of the IGBT is affected by the collector current IC. When IC ≤ 1/3 ICM, the on-resistance RON of IGBT has a negative temperature coefficient, which is not suitable for parallel connection. When IC > 1/3 ICM, the on-resistance RON of IGBT presents a positive temperature coefficient, which is suitable for parallel connection as the same as MOSFET. When the current is small, the impact of uneven current on the IGBT is relatively small, so overall IGBTs are still very suitable for parallel connection.
Through the composite structure, a fully-controlled device that integrates the advantages of multiple devices can be manufactured, such as MCT, SIT, SITH, IGCT, etc.
The MOS controlled thyristor (MCT) combines the advantages of MOSFET and thyristor, which has the advantages of extremely high di/dt and dv/dt tolerance, fast switching speed, low on-state voltage, small switching loss, high voltage capacity, and high current capacity. Similar to GTO, the MCT is composed of tens of thousands of MCT units, and each unit is composed of a PNP thyristor and a MOSFET. The MOSFET controls the working state of the PNP thyristor. However, although the idea of MCT is similar to that of IGBT, its voltage and current capacity are far from the expected value, and the cost is higher than that of IGBT, so it cannot be put into the market.
The static induction transistor (SIT) is a kind of the junction field effect transistor with majority carriers participating in conduction process. The operating frequency of SIT is equivalent to or even higher than that of Power MOSFET, and its power capacity is larger than that of MOSFET, so it is suitable for high frequency and high power applications. SIT is a normally open (NO) switch -- when no signal is applied, the SIT is turned on; when a negative bias is applied, the SIT is turned off. In practical applications, the normally open switch is not as safe as the normally closed switch. In addition, the on-state resistance of SIT is large, and the on-state loss is also large, so SIT cannot be widely used like Power MOSFET. SIT is mainly used in the fields of radar communication equipment, ultrasonic power amplification, pulse power amplification and high-frequency induction heating.
The static induction thyristor (SITH) is a bipolar field controlled thyristor (FCT). Many characteristics of SITH are similar to GTO. SITH has the advantages of conductance modulation effect, low on-state voltage, and large current capacity. The switching speed of SITH is much higher than that of GTO, and the current turn-off gain is smaller than of GTO. SITH has both normally open and normally closed types.
The integrated gate-commutated thyristor (IGCT, or GCT) combines the advantages of IGBT and GTO. Its power capacity is equivalent to GTO, and the switching speed is 10 times that of GTO. IGCT does not need complicated buffer circuit, but its driving power is still very large. Both IGCT and IGBT may replace GTO in the high-power field.
● General Purpose Diode (GPD)
Advantages: High reverse voltage peak voltage, low forward voltage drop, strong rectification ability.
Disadvantages: Long reverse recovery time and low operating frequency.
● Fast Recovery Diode (FRD)
Advantages: Short reverse recovery time, high operating frequency, low forward voltage drop, and high peak reverse voltage.
Disadvantages: Weak rectification capability.
● Schottky Barrier Diode (SBD)
Advantages: Extremely short reverse recovery time, extremely high operating frequency, and extremely low forward voltage drop.
Disadvantages: Normal rectification ability, low peak reverse voltage, high temperature sensitivity, and large leakage current.
● Fast Switching Thyristor (FST)
Advantages: Short switching time, high working frequency, high dv/dt and di/dt tolerance.
Disadvantages: Low rated voltage and low rated current.
● Triode AC Switch (TRIAC)
Advantages: The reverse characteristic is the same as the forward characteristic, and it can work in an AC circuit.
Disadvantages: No reverse blocking capability.
● Reverse Conducting Thyristor (RCT)
Advantages: Low on-state voltage, short turn-off time, operating frequency significantly higher than FST, high rated junction temperature, high voltage capacity, integrated a power diode which can simplify circuit design.
● Light Triggered Thyristor (LTT)
Advantages: Optical triggering can ensure good insulation between the control circuit and the main circuit, strong anti-electromagnetic interference ability, very large current capacity, and very large voltage capacity. It is currently the power electronic device with the highest power capacity.
Disadvantages: The operating frequency is generally not high.
● Gate Turn-off Thyristor (GTO)
Advantages: Large voltage capacity, large current capacity, suitable for high-power applications, with conductance modulation effect, low current turn-off gain, and good thermal stability.
Disadvantages: Low switching speed, large negative gate pulse turn-off current, large driving power, and complex driving circuit.
Remarks: GTO is the first choice for high power (megawatt level), high voltage, and low switching frequency.
● Giant Transistor (GTR)
Advantages: Large voltage capacity, large current capacity, suitable for medium-power applications, good switching characteristics, low saturation voltage drop.
Disadvantages: Low switching speed, current drive, large drive power, complex drive circuit, second breakdown, higher price than IGBT.
Remarks: With the capacity of IGBT increases, GTR will gradually withdraw from the stage of history.
● Power MOSFET
Advantages: Fast switching speed, high input impedance, good thermal stability, low driving power, simple driving circuit, high operating frequency, and no second breakdown.
Disadvantages: Small current capacity, low withstand voltage, generally only suitable for power electronic devices with a power not exceeding 10kW.
Remarks: Power MOSFET is the first choice for small and medium power, low voltage and high switching frequency.
● Insulated Gate Bipolar Transistor (IGBT)
Advantages: High switching speed, low switching loss, ability to withstand pulse current impact, low on-state voltage drop, high input impedance, voltage drive, low drive power, no second breakdown.
Disadvantages: Switching speed is lower than Power MOSFET, voltage and current capacity are not as good as GTO.
Remarks: IGBT is the first choice for medium power, high voltage and low switching frequency.
● MOS Controlled Thyristor (MCT)
Advantages: Able to withstand extremely high di/dt and dv/dt, extremely fast switching speed, low conduction voltage drop, small switching loss, high voltage capacity, and high current capacity.
Disadvantages: The voltage capacity and current capacity are smaller than IGBT, but the cost is higher than IGBT.
● Static Induction Transistor (SIT)
Advantages: The operating frequency is larger than that of Power MOSFET, and the power capacity is larger than that of Power MOSFET.
Disadvantages: Normally open, large on-state resistance, large on-state loss.
● Static Induction Thyristor (SITH)
Advantages: Conductance modulation effect, low on-state voltage drop, large current capacity, switching speed is higher than that of GTO, and current turn-off gain is smaller than that of GTO.
Disadvantages: Mostly normally open.
● Integrated Gate-Commutated Thyristor (IGCT)
Advantages: The power capacity is equivalent to GTO, the switching speed is 10 times that of GTO, and no complicated buffer circuit is required.
Disadvantages: High driving power.
In the previous chapters, it is mentioned that more and more power electronic devices have begun to be modularized. Modularity refers to packaging multiple devices in a module to reduce the manufacturing cost of the device and reduce the volume of the device. At the same time, because modularization reduces the leads in the circuit, the circuit design is simplified, and the circuit inductance is also greatly reduced, thereby reducing the demand for protection circuits and buffer circuits, and further simplifying the circuit design, making the circuit more reliable.
Common power modules include solid-state thyristor modules, solid-state power diode rectifier modules, solid-state fully-controlled bridge rectifier modules, and solid-state half-controlled bridge rectifier modules. You can click the product page to get more information about the power module.
Power electronic devices cannot be used directly, and a power electronic system (PES) needs to be built first. The power electronic system is composed of a main circuit, a control circuit, a drive circuit, a detection circuit and a protection circuit.
Main Circuit: It is used to realize the change or control of electric energy. Power electronic devices are the core components of the main circuit.
Control Circuit: It is used to provide control signals to the drive circuit.
Drive Circuit: It is used to convert the control signal of the control circuit into a gate signal for the main circuit.
Detection Circuit: It is used to detect the working status of the main circuit and feed it back to the control circuit.
Protection Circuit: It is used to protect the control circuit and the main circuit to ensure the reliable operation of the entire system.
Electrical Isolation: It is used to isolate the control circuit (small current) from the main circuit (large current).
As the most common power electronic device, the internal structure of a solid state relay is a basic power electronic system (The working principle of Zero-Crossing AC Solid-State Relays).
The drive circuit is the interface between the main circuit and the control circuit, and is used to convert the control signal of the control circuit into a turn-on signal or a turn-off signal for the main circuit. For half-controlled devices, the drive circuit only needs to provide a turn-on signal. For fully-controlled devices, the drive circuit should provide a turn-on signal and a turn-off signal. When designing the drive circuit, many factors need to be considered to make the power electronic devices work in an ideal state. A good driving circuit can effectively decrease the switching time and switching power consumption of power electronic devices, and at the same time can ensure the working efficiency, safety and reliability of power electronic devices.
According to the type of the drive signal, the drive circuit can be divided into the current-driven circuit and the voltage-driven circuit. The current drive circuit can provide a current drive signal and a threshold voltage for the current-driven device; the voltage drive circuit can provide a voltage drive signal for the voltage-driven device. The voltage-driven circuit is easier to design and manufacture than the current-driven circuit, so the voltage-driven devices are more popular. The drive circuit can also be divided into the discrete drive circuit and the integrated drive circuit. The discrete drive circuit need to be designed separately for different power electronic devices, and also need to consider issues such as parameter matching and electromagnetic compatibility. Therefore, in order to achieve the best performance of the power electronic device, the integrated drive circuit specially developed by the device manufacturer is usually preferred.
Since the control circuit is usually formed by information electronic devices with a very low operating power, and the power of the main circuit is very large, the drive circuit also needs to provide electrical isolation between the control circuit and the main circuit to prevent the signal from the main circuit from damaging the control circuit. The isolation method can be usually divided into optical isolation and transformer isolation (magnetic isolation).
Optical isolation refers to the transmission of the control signal to the main circuit through light. Generally, a photocoupler (optocoupler, OPT) is used for optical isolation, and the light-emitting diode (LED) in the optocoupler controls the phototransistor through optical signals. The input current ID of the photocoupler is equivalent to the base current of a general transistor, and the output current IC is equivalent to the collector current of a general transistor. The input and output characteristics of optocouplers are similar to BJT. The current gain ID/IC of ordinary optocouplers is usually less than 1. The high transmission ratio type optocoupler will use Darlington structure to increase its current gain, but the withstand voltage of the optocoupler is limited (generally within 2000V). The light triggered thyristor transmits the control signal through the optical fiber, so there is no need to add additional optical isolation measures. At the same time, the light triggered thyristor is a half-controlled device, so there is no need to consider how to turn it off.
Transformer isolation refers to the isolation of the primary winding and the secondary winding through the iron core of the pulse transformer, and the use of the magnetic saturation characteristics of the iron core to transmit the control signal from the input circuit to the output circuit. Due to the non-linear distortion characteristics of transformers, pulse transformers are generally suitable for high-frequency signals, so heating and loss must also be considered. Since the pulse transformer generally works at the initial permeability of the core, the volume of the pulse transformer will be much larger than other transformers.
● Ensure a certain pulse width (pulse sustain time) so that the internal positive feedback of the thyristor can be established to ensure the reliable conduction of the thyristor.
● Ensure a certain pulse flat top amplitude to provide sufficient drive current, and at the same time avoid the appearance of burrs to interfere with the normal operation of the thyristor.
● Ensure that the gate voltage, gate current and gate power are all within the rated trigger area to avoid damage to the SCR.
● Provide necessary protective measures, such as electrical isolation, temperature control, anti-interference, etc.
● If the load power is large, a pulse transformer (PTR) can be used as the electrical isolation device, as shown in Figure 61, a. In order to obtain a gate pulse current with a sufficiently large amplitude, a sufficiently long duration, and the shortest possible current rise time, a transistor amplifier (TRA) is usually added to the input side of the pulse transformer (PTR).
● If the load power is small, a optocoupler (OPT) can be used as an electrical isolation device, as shown in Figure 61, b.
The parameters of fully-controlled power electronic devices are different, so their drive circuits are also different. Generally, the current drive type drive circuit is usually more complicated than the voltage drive type. It is generally recommended to directly use the integrated drive circuit provided by the manufacturer to ensure the performance of power electronic devices as much as possible.
● Same as that of thyristor.
● It is necessary to provide a turn-off current (provided by the gate reverse bias circuit) much larger than the turn-on current to speed up the turn-off process of GTO.
The drive circuit of GTO usually includes turn-on drive circuit, turn-off drive circuit and gate reverse bias circuit. And the drive circuit of GTO can be divided into the pulse transformer coupling type and the direct coupling type. The common discrete drive circuit of GTO is shown in Figure 62:
● Pulse transformer coupling drive circuit is shown in Figure 62, a. The pulse transformer provides electrical isolation. And the pulse signal on the secondary winding of the pulse transformer directly drives the GTO. However, the pulse transformer has leakage inductance, which will affect the switching speed of the GTO, and also lead to the inability to form a steep pulse front. The direct coupling circuit can avoid this problem.
● Direct coupling drive circuit is shown in Figure 62, b. The isolation circuit (transformer or optocoupler) provides electrical isolation; VD1 and C1 provide +5V voltage; VD2, VD3, C2, C3 form a voltage doubling rectifier circuit, providing +15V voltage; VD4 and C4, provide -15V voltage. When V1 is turned on, it outputs a strong positive pulse. When V2 is turned on, it output a flat top part of the positive pulse. When V2 is turned off and V3 is turned on, it outputs the negative pulse. When V3 is turned off, R3 and R4 provide negative gate bias voltage. The direct coupling drive circuit can avoid mutual interference and parasitic oscillation within the circuit, and can provide a relatively steep pulse front, but its design is complicated, power consumption is large, and efficiency is low. However, considering the comprehensive factors, the direct coupling drive circuit still has a wide range of applications.
● Same as that of thyristor.
● A certain negative base current needs to be applied to reduce the turn-off time and turn-off loss.
● A negative bias of a certain amplitude (about 6V) needs to be applied between the base and the emitter to make the GTR turn-off more reliable.
The common discrete drive circuit of GTR is shown in Figure 63. The transistor amplifier circuit is between the main circuit and the optocoupler, which is mainly composed of a transistor amplifier. The main circuit is at a low potential when it is turned on, and at a high potential when it is turned off. Therefore, the transistor amplifier circuit needs to be powered separately and electrically isolated from the main circuit power supply to avoid potential fluctuations of the main circuit from affecting the working of the amplifier circuit.
When the load is small, if all the emitter current of V4 is injected into the GTR, it will oversaturate the GTR, and its de-saturation time will be prolonged when it is turned off. Therefore, the clamp power diode VD2 and the potential compensation power diode VD3 are usually added to form a Baker clamp anti-saturation circuit. When the GTR is oversaturated, the collector potential is lower than the base potential, VD2 will automatically turn on, so that the excess drive current flows into the collector, maintaining Ubc≈0. With the help of Baker circuit, the GTR is in critical saturation when it is turned on, and it is easy to be turned off.
The accelerating capacitor C2 is used to speed up the switching process of GTR. When the GTR is turned on, R4 is short-circuited by C2, which causes the drive current to overshoot and at the same time increases the steepness of the pulse front to speed up the turn-on process. When V4 is turned on, C2 will be charged to prepare for the turn off of GTR, and its charging polarity is left positive and right negative. When V4 is off and V5 is on, the charging voltage on C2 provides a reverse voltage for the emitter junction of the GTR, so that the GTR will be quickly turned off.
The forward characteristic of the unidirectional transient voltage suppression power diode TVS is the same as that of the ordinary stabilized voltage power diode, and its reverse characteristic is a typical PN junction avalanche device. TVS has the advantages of fast response time, large transient power, low leakage current, small size, and easy to control the clamping voltage. TVS can also effectively absorb surge pulses and eliminate crosstalk. When a transient overvoltage pulse is applied to both ends of the TVS, its impedance will change from high impedance to low impedance within a few picoseconds to absorb up to several kilowatts of surge power and clamp its voltage at both ends to the preset value.
● Need to provide a stable turn-on drive voltage: For MOSFET is generally 10-15V; for IGBT is generally 15-20V.
● The output resistance of the drive circuit needs to be small enough (tens to hundreds of ohms) to quickly establish the drive voltage.
● A resistor with low resistance (tens of ohms) needs to be connected in series to the gate to consume feedback energy, reduce the gain of the amplifier circuit, and reduce parasitic oscillation.
● It is necessary to apply a certain amplitude of negative driving voltage (usually 5-15V) to reduce the turn-off time and turn-off loss, and improve the reliability of the turn-off.
Similar to the drive circuit of GTR, the voltage drive circuit also includes an electrical isolation circuit and a transistor amplifying circuit, as shown in Figure 64. However, the voltage drive circuit only needs to provide a voltage drive signal, so its design is much simpler than that of GTR.
Parasitic oscillation refers to the oscillation that is generated by the internal parasitic parameters of the transistor amplifier, which has nothing to do with the operating frequency or is not within the operating frequency range. Parasitic oscillation can be divided into low-frequency parasitic oscillation (lower than the operating frequency) and high-frequency parasitic oscillation (higher than the operating frequency). Even if the input terminal of the transistor amplifier is short-circuited, there is usually an oscillating signal at the output terminal. But if handled properly, the parasitic oscillation can be completely eliminated. Parasitic oscillation has the following characteristics:
● The oscillation period of the parasitic oscillation is generally regular and the waveform is relatively regular.
● The amplitude of parasitic oscillation is generally large, and sometimes it can even trigger the turn-on and turn-off of power electronic devices.
● Except for low frequency oscillation caused by poor power supply decoupling, the oscillation frequency of parasitic oscillation is generally high, and the oscillation frequency and amplitude will vary with the component parameters of the transistor amplifier.
The protection circuit can provide a safe and reliable working environment for power electronic devices. The reasons for the damage of power electronic devices are mainly divided into overcurrent and overvoltage. In the event of overcurrent, the internal structure of power electronic devices will be destroyed due to a sharp rise in temperature. Overvoltage is usually accompanied by overcurrent, which will also cause damage to the internal structure of power electronic devices. Due to the complexity of the circuit and the consideration of safety, there is usually no single protection measure is taken, but as much protection as possible for the circuit within the cost range.
1.1- External Overvoltage (mainly from lightning strikes and system operation process)
● Lightning Overvoltage: The overvoltage caused by lightning.
● Operating Overvoltage: During the opening and closing of the upper circuit, the overvoltage caused by the leakage inductance of the transformer in the power grid.
1.2- Internal Overvoltage (mainly comes from the switching process of power electronic devices)
● Turn-off Overvoltage: When the power electronic device is turned off, the circuit inductance induces an overvoltage at both ends of the device due to the rapid decrease of the forward current.
● Commutation Overvoltage: After the commutation of the bidirectional thyristor or anti-parallel power diode, its reverse current decreases sharply, so the line inductance will generate overvoltage at both ends of the device. Commutation means that a bidirectional device with a center symmetrical output characteristics switches back and forth between the working states of the first quadrant and the third quadrant. The commutation process inevitably accompanies the turn-on and turn-off of power electronic devices, so there must be a turn-off overvoltage during the commutation process.
● Lightning Arrester F: It is used to conduct lightning overvoltage to the ground. The lightning arrester is generally a zinc oxide arrester, which has the same characteristics as a varistor.
● Electrostatic Shielding Layer of the Transformer D: It is used to conduct high-voltage static electricity to the ground.
● Suppression capacitor C: It is used to suppress overvoltage. However, the voltage absorbed by the capacitor will generate a larger discharge current during the discharge process.
● Varistor RV: The resistance value of the varistor decreases as the voltage increases, so It conduct the overvoltage to other circuit or the ground.
● RC1 Protection Circuit: The RC circuit adds a resistor R on the basis of the suppression capacitor C. Although its ability to absorb the overvoltage is poor, it can effectively suppress the discharge current during the discharge of the capacitor.
● RC2 Protection Circuit: The overvoltage of the external circuit will inevitably produce the DC voltage of the RC2 circuit, so the capacitor C needs to have two discharge circuits to absorb these two voltages through the resistance to leave enough margin value to deal with the next overvoltage.
● RC3 protection circuit: It is used to suppress the commutation overvoltage.
● RCD protection circuit: It is used to suppress the turn-off overvoltage.
1.1- Overload Overcurrent: It refers to that the stable working current exceeds 120% of the rated value.
1.2- Short-circuit overcurrent: It refers to a current that has a short duration and is several times larger than the rated current.
● Current Transformer CT: It is a special transformer for measurement. It consists of a closed iron core column and two mutually insulated coils sheathed on the iron core. The coil connected to the measured line is the primary side of CT. The coil connected to the detection circuit is the secondary side of CT. The primary side and the secondary side of CT are electrically isolated from each other. The main function of CT is to convert the current of the circuit under test into the current range that the detection circuit can detect. When using a current transformer, it is necessary to avoid an open circuit on the secondary side of the current transformer, because an open circuit on the secondary side will make the current transformer lose the demagnetization effect of the secondary winding. Therefore, the current of the primary winding all forms the excitation current, which increases the magnetic flux in the iron core to oversaturate the iron, which eventually leads to heat and damage to the current transformer. At the same time, if the secondary winding has a large number of turns, high voltage will be induced, which will endanger the safety of operators and equipment.
● Over-current Relay KA: When the overcurrent in the circuit is detected, the overcurrent relay starts to act, causing the AC circuit breaker QF1 to open. The action process of the overcurrent relay is relatively long, but it can directly cut off the power source of the equipment to avoid subsequent damage to the operator and the equipment.
● DC Fast Circuit Breaker QF2: The DC fast circuit breaker uses the strong short-circuit electric repulsion force to directly push the contacts open with very short action time (2-3 milliseconds), so that the short-circuit current will be shut off before reaches its maximum value. The DC fast circuit breaker is also equipped with an arc extinguishing chamber, so it has good current limiting performance.
● Fast Fuse FU: The protection methods of fast fuse can be divided into short-circuit protection and full protection (overload protection and short-circuit protection). Fast fuse is not suitable for protecting high frequency devices (such as IGBT and MOSFET). Fast fuse is usually only suitable for protecting thyristor, because the thyristor has strong current withstand capability, and both thyristor and fast fuse have second-ampere characteristics (the relationship between the overcurrent value and the maximum overcurrent withstand time). So it is easy to select the corresponding fast fuse based on the second-ampere characteristics of thyristor to ensure that the fast fuse shuts off the output circuit of the thyristor before the thyristor is damaged. Fast fuse is a disposable device, if it is blown, the fast fuse must be replaced with the same specification.
● Electronic Protection Circuit: The electronic protection circuit is mainly used to turn off the trigger circuit to make the on-state fully-controlled device switch to the high-impedance off-state, so that other power electronics devices will not be turned on and maintain the off-state. This method is invalid for the half-controlled device that has been turned on. Compared with fast fuse, the cost of electronic protection circuit is very low and the action speed is very fast, so the electronic protection circuit can effectively protect medium and high frequency power electronic devices. The electronic protection circuit is generally designed directly into the drive circuit.
The snubber circuit (also known as the absorption circuit, or buffer circuit) suppresses the current rise rate (di/dt) of the power electronic device by using the characteristic that the inductor current cannot change suddenly, and suppresses the voltage rise rate (dv/dt) of the power electronic device by using the characteristic that the capacitor voltage cannot change suddenly. The most basic RLCD buffer circuit is shown in Figure 67, a. When the device is turned on, the inductance L suppresses di/dt; when the device is turned off, the capacitor C is charged through the fast power diode VD to absorb the overvoltage that appears on the device and limit the re-applied dv/dt. When the device is turned on, the energy on C is consumed by the resistor R. In order to reduce power loss of the device with high operating frequency and small capacity, the RLCD snubber circuit can be simplified to the RCD snubber circuit, as shown in Figure 67, b.
According to the destination of energy, the snubber circuit can be divided into an energy-consuming snubber circuit and an energy-feeding snubber circuit. The energy-consuming snubber consumes excess energy with resistors. The energy-feeding snubber circuit has no resistor, and will use the excess energy.
According to the application, the snubber circuit can be divided into turn-on snubber circuit (di/dt suppression circuit), turn-off snubber circuit (dv/dt suppression circuit) and composite snubber circuit (combination of turn-on snubber circuit and turn-off snubber circuit). The turn-on snubber circuit is used to suppress the current overshoot and di/dt to reduce the turn-on loss of the power electronic device. The turn-off snubber circuit is used to absorb the overvoltage generated by the line inductance and suppress the dv/dt to reduce the turn-off loss of the power electronic device. The current power electronic device has a high di/dt tolerance and high current tolerance, so usually there is no need to add an additional turn-on buffer circuit.
The fully controlled self-shut-off device (such as GTR and GTO) must use composite snubber circuits. The turn-on snubber circuit of GTR is used to suppress the di/dt during the turn-on process, so as to avoid overheat and second breakdown. It also plays a role in suppressing the di/dt and the peak penetration short-circuit current in the GTR inverter. The turn-off snubber circuit of GTO is not only to limit the dv/dt and overvoltage of the re-applied voltage when the GTO is turned off, but also to reduce the turn-off loss of the GTO, so that the GTO play its due turn-off capability and give full play to its load capacity. The function of the snubber circuit of IGBT focuses more on the absorption and suppression of overvoltage during the switching process. This is because the operating frequency of the IGBT is very high, so even a small circuit inductance produces a high di/dt, which leads to overvoltage and endangers the safety of the IGBT. The Power MOSFET uses a snubber circuit to suppress di/dt and dv/dt, mainly to change the switching process trajectory of the device, to reduce the switching loss, and to make the device operate reliably.
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